WO2016181513A1 - Power supply device - Google Patents

Abstract

[Problem] To provide a technology of improving efficiency of a power supply device that supplies a load with direct current power. [Solution] A control circuit (140) turns on a switching element (T131) in a set cycle. Consequently, a capacitor (C132) discharges electricity via a parallel circuit of an inductor (L131), a capacitor (C131), and a load (120). Furthermore, the control circuit (140) turns off the switching element (T131) when a current flowing in the switching element (T131) becomes higher than a threshold interlocking with the magnitude of a direct current voltage. Consequently, a reflux current flows via a diode (D131) and a parallel circuit of the inductor (L131), the capacitor (C131), and the load (120). Furthermore, while the switching element (T131) is turned off, the capacitor (C132) is charged via an inductor (L132). The inductance of the inductor (L132) is set to a value larger than the inductance of the inductor (L131).

Description

Power supply

The present invention relates to a power supply device that supplies DC power to a load, and more particularly to a power supply device that can improve power factor and efficiency.

Conventionally, a power supply device disclosed in Japanese Patent Laid-Open No. 9-47024 is known as a power supply device that supplies DC power to a load. FIG. 10 shows a circuit diagram of a conventional power supply device 800.
The power supply device 800 shown in FIG. 10 is configured as a step-down AC-DC converter including a full-wave rectifier circuit 810, a low-pass filter 900, a power supply circuit 830 that supplies DC power to a load 820, and a control circuit 840. Yes. Full-wave rectification circuit 810 performs full-wave rectification on AC voltage Vac of AC power supply 10 and converts it into DC voltage (pulsating DC voltage) Vdc. An N-type MOSFET is used as the power switching element T831 of the power supply circuit 830. In addition, the power supply circuit 830 has a power factor correction (PFC) function for suppressing the amplitude of the harmonic current below a limit value (referred to as “PFC standard”). The control circuit 840 turns on the switching element T831 in the set cycle, and turns off the switching element T831 when the drain current of the switching element T831 (voltage drop of the current detection resistor R831) becomes larger than the threshold value.

In power supply device 800 shown in FIG. 10, when switching element T831 is turned on, full-wave rectifier circuit 810, low-pass filter 900, inductor L831, a parallel circuit of load 820 and capacitor C831, switching element T831, current detection resistor R831, and A current I1 flows through a ground path. At this time, electromagnetic energy is accumulated in the inductor L831 (inductance L). When the switching element T831 is turned off, the electromagnetic energy accumulated in the inductor L831 causes the inductor L831, the parallel circuit of the load 820 and the capacitor C831, and the path of the diode D831 (referred to as “freewheel diode” or “freewheel diode”). A reflux current I3 flows.

In the power supply device 800 shown in FIG. 10, the magnitude (amplitude) of the current I1 is linked to the magnitude (amplitude) of the DC voltage Vdc. As a result, the power factor cos θ (θ: phase difference between the AC voltage Vac and the AC input current Iac) of the AC input power approaches “1”.
The current I1 is cut off while the switching element T831 is off. That is, the current I1 flows intermittently. For this reason, the harmonic current is included in the current I1. A low-pass filter 900 is provided in order to prevent this harmonic current from propagating to the AC power supply side.

JP-A-9-47024

The low-pass filter 900 provided in the conventional power supply device 800 includes a capacitor having a small capacitance and an inductor having a large inductance in order to prevent the power factor from deteriorating. An inductor having a large inductance has a large number of turns of the conductive wire. Here, when an inductor is formed using a thick conducting wire, the inductor becomes large, and thus it is necessary to form the inductor using a thin conducting wire. On the other hand, when an inductor is formed using a thin conducting wire, the resistance value of the inductor increases and the loss of the inductor increases.
As described above, the conventional power supply apparatus has a limit in improving the efficiency because the loss of the inductor constituting the low-pass filter for preventing the harmonic current from propagating to the AC power supply side is large.
The present invention has been made in view of such a point, and an object of the present invention is to provide a power supply device capable of improving the power factor and improving the efficiency.

The present invention includes a rectifier circuit that converts an AC voltage from an AC power source into a DC voltage, a power supply circuit that supplies DC power to a load, and a control circuit that controls the power supply circuit.
The rectifier circuit converts an AC voltage into a DC voltage (pulsating DC voltage), and generates the converted DC voltage between the positive electrode end and the negative electrode end. As the rectifier circuit, a full-wave rectifier circuit is typically used. The terms “positive electrode end” and “negative electrode end” are used as terms representing locations where a DC voltage is generated.
The power supply circuit includes first and second capacitors, first and second inductors, diodes, and switching elements.
In the first invention, the second inductor and the second capacitor are arranged in series between the positive terminal and the negative terminal of the rectifier circuit. Between the terminals of the second capacitor, a first inductor, a parallel circuit of a load and a first capacitor, and a switching element are arranged in series. A diode (freewheeling diode) is arranged in parallel in the series circuit of the first inductor and load and the parallel circuit of the first capacitor.
The inductance of the second inductor is set to a value larger than the inductance of the first inductor.
The control circuit is configured to turn on the switching element at a set cycle, and to turn off the switching element when the current flowing through the switching element becomes greater than a threshold linked to the magnitude of the DC voltage. As the “threshold value linked to the magnitude of the DC voltage”, a “threshold value proportional to the magnitude (amplitude) of the DC voltage” is typically used.
Thereby, when the switching element is turned off, the second capacitor is charged by the DC voltage of the rectifier circuit. Further, when the switching element is turned on, the second capacitor is discharged through a path including the first inductor, the load, and the switching element. When the switching element is turned off, the electromagnetic energy accumulated in the first inductor when the switching element is turned on is converted into a return current that flows through a path including the first inductor, the load, and the diode.
In the second invention, the second inductor, the first inductor, and the second capacitor are arranged in series between the positive terminal and the negative terminal of the rectifier circuit. A parallel circuit of a load and a first capacitor and a switching element are arranged in series at both ends of the series circuit of the first inductor and the second capacitor. A diode (freewheeling diode) is arranged in parallel in the series circuit of the first inductor and load and the parallel circuit of the first capacitor.
The inductance of the second inductor is set to a value larger than the inductance of the first inductor.
The control circuit is configured to turn on the switching element at a set cycle and turn off the switching element when the current flowing through the switching element becomes greater than a threshold linked to the magnitude of the DC voltage, as in the first invention. ing.
Thereby, when the switching element is turned off, the second capacitor is charged by the DC voltage of the rectifier circuit. Further, when the switching element is turned on, the second capacitor is discharged through a path including the first inductor, the load, and the switching element. When the switching element is turned off, the electromagnetic energy accumulated in the first inductor when the switching element is turned on is converted into a return current that flows through a path including the first inductor, the load, and the diode.
In the first and second inventions, fluctuations in the AC input current flowing during each control cycle (ON period + OFF period) of the switching element can be suppressed, and the AC input current can be synchronized with the AC voltage. . As a result, the generation of harmonic current can be suppressed, the low-pass filter for preventing the harmonic current from propagating to the AC side can be removed, or the inductance of the inductor constituting the low-pass filter can be reduced. Can be reduced. That is, the power supply device of the present invention can improve the power factor and the efficiency.
In a different form of the first invention, the control device is configured to execute the processing every clock cycle. When the DC voltage is in the increasing process, a value obtained by subtracting a value corresponding to the charge amount of the second capacitor due to the increase in the DC voltage during the clock period from the threshold is used as the threshold, and the DC voltage is in the decreasing process. In some cases, the threshold value is a value obtained by adding a value corresponding to the discharge charge amount of the second capacitor due to the decrease in the DC voltage during the clock cycle.
In this embodiment, since the generation of the harmonic current can be further suppressed by increasing the capacity of the second capacitor, it is possible to remove the low-pass filter for preventing the harmonic current from propagating to the AC side. it can. Thereby, the power supply device of this invention can improve a power factor while improving a power factor more.
In another different form of the first invention, the threshold value is configured to be linked to a value obtained by subtracting the set value from the magnitude of the DC voltage. Then, when the magnitude of the DC voltage is smaller than the set value, the control device holds the switching element in the off state.
In this embodiment, since the switching element is controlled so that the DC voltage (pulsating DC voltage) does not fall below the set value, the charge charged in the second capacitor when the DC voltage increases and the switching when the DC voltage decreases. The sum of absolute values of electric charges discharged from the second capacitor when the element is turned on is reduced, and the phase advance by the second capacitor is suppressed. Then, the capacity of the second capacitor can be increased accordingly. Thereby, since generation | occurrence | production of a harmonic current can be suppressed more, the low pass filter for preventing a harmonic current from propagating to the alternating current side can be removed.
In another different form of the first and second inventions, a third capacitor that can be arranged in series with the second capacitor is provided. Preferably, a changeover switch for connecting the third capacitor in series with the second capacitor or for disconnecting the connection between the third capacitor and the second capacitor is used.
For example, if the DC power supplied to the load is decreased by changing the threshold value in a state where the capacity of the second capacitor is set to a predetermined value, the power factor may deteriorate. In the present embodiment, in such a case, the power factor can be prevented from deteriorating by connecting the third capacitor in series with the second capacitor to reduce the combined capacitance.
In another different form of the first and second inventions, a MOSFET parasitic diode is used as the diode (freewheeling diode). Then, the control circuit turns on the MOSFET within a set period after the switching element is turned off and when the cathode voltage of the parasitic diode becomes smaller than a value obtained by subtracting the first set value from the anode voltage, and the set period elapses. Later, when the cathode voltage of the parasitic diode becomes larger than the value obtained by subtracting the first set value from the anode voltage, the MOSFET is turned off. The setting period is set to a period shorter than the period from when the return current starts to flow through the parasitic diode of the MOSFET until it disappears.
In this embodiment, when the return current is passed through the MOSFET's parasitic diode, the MOSFET is turned on, so that the power loss is reduced and the efficiency is reduced as compared with the case where the return current is passed through the return diode. Can be improved.
In the case where the reflux current is passed through the parasitic diode of the MOSFET, it is preferable to use a voltage detection circuit that detects the voltage between the anode and the cathode of the parasitic diode in order to determine the on / off timing of the MOSFET.
In another different form of the first and second inventions, a voltage detection circuit having first and second source follower circuits constituted by FETs is used.
The source resistance of the first source follower circuit is connected to the anode of the parasitic diode, and the source resistance of the second source follower circuit is connected to the cathode of the parasitic diode. A common bias voltage is applied to the gates of the FETs of the first and second source follower circuits.
In this embodiment, the voltage between the anode and the cathode of the parasitic diode is converted into a difference between the current flowing through the first source follower circuit and the current flowing through the second source follower circuit.
In another different form of the first and second inventions, a voltage detection circuit having first and second circuits is used.
The first circuit includes a diode arranged in series between the cathode of the parasitic diode and the ground, a first resistor, a first PMOSFET, a second resistor, and a first NMOSFET. The gate and drain of the first PMOSFET are short-circuited, and the gate and drain of the first NMOSFET are short-circuited. The second circuit includes a diode arranged in series between the anode of the parasitic diode and the ground, a third resistor, a second PMOSFET, a fourth resistor, and a second NMOSFET. The gate of the second PMOSFET is connected to the gate of the first PMOSFET, and the gate of the second NMOSFET is connected to the gate of the first NMOSFET. The characteristics of the second PMOSFET and the second NMOSFET have the same characteristics as the first PMOSFET and the first NMOSFET, respectively. The resistance value of the fourth resistor is set to be smaller than the resistance value of the second resistor. The resistance value of the third resistor is set to be equal to or greater than the resistance value of the first resistor.
In this embodiment, the drain voltage of the second NMOSFET is configured to change according to the voltage between the anode and the cathode of the parasitic diode.

The power supply device of the present invention can improve the power factor and the efficiency.
Other features, functions, and advantages of the present invention can be readily understood with reference to the specification, claims, and accompanying drawings.

It is a circuit diagram of a 1st embodiment of a power supply device of the present invention. It is a figure which shows the simulation result at the time of using a threshold value correction circuit. It is a circuit diagram of 2nd Embodiment of the power supply device of this invention. It is a circuit diagram of 3rd Embodiment of the power supply device of this invention. It is a circuit diagram of 4th Embodiment of the power supply device of this invention. It is a circuit diagram of a 5th embodiment of a power supply device of the present invention. It is a figure which shows the simulation result of the power supply device of 5th Embodiment. It is a circuit diagram of 6th Embodiment of the power supply device of this invention. It is a circuit diagram of a seventh embodiment of the power supply device of the present invention. It is a circuit diagram of the conventional power supply device.

The following detailed description is merely to provide those skilled in the art with detailed information for implementing the preferred embodiments of the present invention, and the technical scope of the present invention is not limited by the detailed description. Determined based on the description. For this reason, combinations of configurations and methods in the following detailed description are not all essential to implement the present invention in a broad sense, but in the detailed description described with reference numerals in the accompanying drawings. However, the present invention only discloses typical embodiments of the present invention.

Embodiments of the present invention will be described below with reference to the drawings.
Hereinafter, a case will be described in which a full-wave rectifier circuit that converts an AC voltage into a DC voltage is provided and configured as a step-down AC-DC converter that supplies DC power to a load having a light emitting diode (LED). Of course, the power supply device of the present invention can be used as a power supply device that supplies DC power to various loads other than LED loads.
Further, the terms “voltage” and “current” are used to mean “voltage magnitude (amplitude)” and “current magnitude (amplitude)”, respectively, unless otherwise specified.

[First Embodiment]
FIG. 1 shows a circuit diagram of a first embodiment 100 of the power supply device of the present invention. The power supply device 100 according to the first embodiment includes a full-wave rectifier circuit 110 that converts an AC voltage Vac from an AC power supply 10 into a DC voltage Vdc, a power supply circuit 130, a control circuit 140, a power supply circuit 180, and a drive signal output circuit 190. It is comprised by. The control circuit 140 has a threshold correction circuit 150.
The full-wave rectifier circuit 110 has a plurality of diodes D111 to D114 connected in a bridge, full-wave rectifies the AC voltage Vac applied between the AC input terminals a and b, and between the positive terminal c and the negative terminal d. A DC voltage Vdc is generated. The DC voltage Vdc is a pulsating DC voltage having a magnitude (amplitude) that changes in accordance with the magnitude (amplitude) of the AC voltage Vac.
In the present embodiment, the negative terminal d of the full wave rectifier circuit 110 is grounded. For this reason, grounding any of the terminals means connecting the terminal to the negative electrode end d.

The power supply circuit 130 is disposed between the positive end c and the negative end d of the full-wave rectifier circuit 110 and one end 131 and the other end 132 of the load 120, and supplies DC power to the load 120. The configuration of the power supply circuit 130 will be described below.
A capacitor C131 having a capacitance C1 is disposed between one end 131 and the other end 132 of the load 120.
Between the one end 131 and the positive end c of the load 120, an inductor L131 having an inductance L1 and an inductor L132 having an inductance L2 are arranged in series. At this time, the inductor L131 is connected to one end 131, and the inductor L132 is connected to the positive end c.

A switching element T131 and a current detection resistor R131 are arranged in series between the other end 132 of the load 120 and the ground. In the present embodiment, an N-type MOSFET is used as the switching element T131.
A capacitor C132 having a capacitance Cp is arranged between a connection point r between the inductor L131 and the inductor L132 and a fixed contact e of the changeover switch SW131. The normally closed contact f of the changeover switch SW131 is grounded, and the normally open contact g is grounded via a capacitor C133 having a capacitance Cp1.
A diode D131 is arranged between the other end 132 of the load 120 and the connection point r. At this time, the anode of the diode D131 is connected to the other end 132, and the cathode is connected to the connection point r.

The capacitor C131 corresponds to the “first capacitor” of the present invention, the capacitor C132 corresponds to the “second capacitor” of the present invention, and the capacitor C133 corresponds to the “third capacitor” of the present invention. The inductor L131 corresponds to the “first inductor” of the present invention, and the inductor L132 corresponds to the “second inductor” of the present invention.

The power supply circuit 180 is disposed between the positive terminal c and the ground, and includes a resistor R181, a Zener diode ZD181, and a capacitor C181. The power supply circuit 180 applies a voltage Vzd determined by the Zener voltage of the Zener diode ZD181 to the control circuit 140 and the drive signal output circuit 190.
The drive signal output circuit 190 is disposed between the voltage Vzd and the ground, and includes a switch SW191 and resistors R191 and R192. When the switch SW191 is turned on, the drive signal output circuit 190 outputs an H level drive signal that instructs the start of supply of DC power to the load 120 to the control circuit 140.

The control circuit 140 includes P-type MOSFETs T141, T142, T146, and T147, N-type MOSFETs T143 to T145, T148, and T149, resistors R141 to R144, a constant current source IDC141, a clock signal generation circuit 141, and a Schmitt trigger ST141. , D flip-flop DFF 141, AND circuit AND 141, drive circuit 142, and threshold correction circuit 150.

The magnitude of the current Iref flowing through the series circuit of the resistors R141 and T149 is substantially proportional to the magnitude (amplitude) of the DC voltage Vdc. T148 and T149 and T146 and T147 are arranged to form a current mirror circuit. Let the drain current of T146 be I14. A threshold value generation circuit for generating a threshold value (current threshold value) I14 is configured by T146 to T149 and the resistor R141.
The constant current I11 from the constant current source IDC 141 flows through T145 and the resistor R144, whereby the gate of T145 becomes a constant voltage. The gate voltage at T145 is used as a bias voltage. A first source follower circuit is formed by T144 and the resistor R143, and a current I12 flows through T142 and the first source follower circuit. A second source follower circuit is formed by T143 and the resistor R142, and the current I13 flows through the T141, the second source follower circuit, and the current detection resistor R131.
N-type MOSFETs having the same characteristics are used as T143, T144, and T145. Further, resistors having the same resistance value are used as the resistors R142, R143, and R144. Further, the resistance value of the resistor R142 is set to a value at which the resistance value of the current detection resistor R131 can be ignored compared to the resistance value of the resistor R142.

The drive circuit 142 includes a T-type MOSFET T142a and an N-type MOSFET 142b, and an output terminal is connected to the gate of the switching element T131.
A clock signal is input from the clock signal generation circuit 141 to the reset terminal R bar of the DFF 141 at a set cycle (clock cycle). The voltage at the connection point s between T141 and T143 is input to the clock terminal CLK of the DFF 141 via ST141.

The control circuit 140 turns on the switching element T131 at a set cycle (clock cycle). Further, when the current flowing through the switching element T131 (in this embodiment, the voltage drop of the current detection resistor R131) becomes larger than a threshold value linked to the magnitude (amplitude) of the DC voltage Vdc, the switching element T131 is turned off.
In the present embodiment, the threshold value generated from the threshold value generation circuit is corrected by the threshold value correction circuit 150. The threshold correction circuit 150 will be described later.

Next, the operation of the present embodiment 100 will be described. The resistance value of the current detection resistor R131 is R1, and the resistance value of the resistor R142 is R2.
First, in order to explain the basic operation, the operation of the circuit excluding the threshold correction circuit 150 will be explained.
First, a case where the switch SW191 is off will be described.
At this time, a DC voltage Vdc is generated between the positive terminal c and the negative terminal d of the full-wave rectifier circuit 110. The power supply circuit 180 applies the voltage Vzd to the control circuit 140 and the drive signal output circuit 190.
Thereby, in the control circuit 140, currents I12, I13, Iref and I14 flow, and a threshold value (current threshold value) is set. Here, when the current I14 flows through the resistor R143 and becomes [I12 = I11−I14], the voltage drop of the current detection resistor R131 is “0”, so that [I12 <I11] and the drain of the T141 The current becomes equal to the current I12. Although the drain current of T143 is also equal to the drain current of T141, since the drain current of T143 can flow up to a current having the same magnitude as the current I11, the voltage at the point s is lowered. That is, the voltage at point s is L level.
At this time, since the switch SW191 is off, the drive signal output circuit 190 does not output an H level drive signal.
Therefore, even if the clock signal output from the clock signal output circuit 141 becomes L level and the DFF 141 is reset and the inverted output −Q becomes H level, the output of the AND 141 does not become H level. Therefore, the switching element T131 maintains an off state.
When the switching element T131 is off, the DC voltage Vdc causes the charging current I1 of the capacitor C132 to flow through the inductor L132, the capacitor C132, the fixed contact e of the changeover switch SW131, the normally closed contact f, and the ground path.

When the switch SW191 is turned on, an H level drive signal is output from the drive signal output circuit 190. When the clock signal becomes L level, the DFF 141 is reset and the inverted output −Q becomes H level. In this state, when the clock signal becomes H level next time, the output of the AND 141 becomes H level. As a result, the drive voltage is applied from the drive circuit 142 to the gate of the switching element T131, and the switching element T131 is turned on.
When the switching element T131 is turned on, the capacitor C132, the inductor L131, the parallel circuit of the load 120 and the capacitor C131, the switching element T131, the current detection resistor R131, the ground and the path of the fixed contact e and the normally closed contact f of the changeover switch SW131, A discharge current I2 of C132 flows. At this time, electromagnetic energy ^{[L1 × (I2) 2/} 2] is stored in the inductor L131.

When the current I2 flows through the current detection resistor R131 (resistance value R1), a voltage drop [R1 × I2] occurs. As a result, the current value that T143 can flow is reduced by [R1 × I2 / R2]. When I2 increases and [(R1 × I2 / R2)> I14], the current value that T143 can flow is less than I12, the voltage at point s rises, and the output of ST141 becomes H level. As a result, an H level signal is input to the clock terminal CLK of the DFF 141, and the inverted output -Q becomes L level. Therefore, the switching element T131 is turned off.
When the current I2 becomes larger than [(R2 / R1) × I14], the switching element T131 is turned off. Therefore, [(R2 / R1) × I14] is a threshold value for controlling the switching element T131 based on the current I2. Become.

Since the current I14 is proportional to the magnitude (amplitude) of the DC voltage Vdc, the threshold value changes when the DC voltage Vdc changes. The threshold for the same DC voltage Vdc can be changed by changing the proportionality constant between the DC voltage Vdc and the current I14. As a method of changing the proportionality constant, a method of changing the resistance value of the resistor R141, or a method of changing the current mirror ratio of T148 and T149 or T146 and T147 can be used.

When the switching element T131 is turned off, the return current I3 flows through the inductor L131, the parallel circuit of the load 120 and the capacitor C131, and the path of the diode D131 due to the electromagnetic energy accumulated in the inductor L131. Hereinafter, a diode through which a return current flows is referred to as a “return diode”.
At the same time, the charging current I1 of the capacitor C132 flows through the path of the fixed contact e and the normally closed contact f of the full-wave rectifier circuit 110, the inductor L132, the capacitor C132 and the changeover switch SW131. Thereby, the charge of the capacitor C132 discharged when the switching element T131 is turned on is supplemented.
At this time, if the inductance L2 of the inductor L132 is small, charging of the capacitor C132 is completed during the OFF period of the switching element T131, and the current I1 becomes “0”. On the other hand, when the inductance L2 of the inductor L132 is increased, the charging current I1 continues to flow during the OFF period of the switching element T131.
Here, the current I1 is an absolute value of the AC input current Iac. Therefore, by continuing the current I1 during the OFF period of the switching element T131, the AC input current Iac continues to flow during one cycle (ON period + OFF period) (also referred to as “control period”) of the switching element T131. Thus, the generation of harmonic current is suppressed.
As a result, the low-pass filter used in the conventional power supply device 800 shown in FIG. 10 can be removed. Alternatively, even when the low pass filter cannot be removed, the inductance of the inductor constituting the low pass filter can be reduced. Therefore, the efficiency can be improved and the product shape can be greatly reduced.

The charging / discharging operation of the capacitor C132 will be described below.
When the proportionality constant between Vdc and I14 is K, I14 is expressed as [I14 = K × Vdc].
If Is is a threshold value for blocking I2 (turning off the switching element T131), Is is expressed by (Equation 1).
Is = (R2 / R1) × I14 = (R2 / R1) × K × Vdc (Formula 1)
Since the resistance value of the path through which I2 flows is very small, when switching element T131 is turned on, I2 increases linearly.

When the current (load current) of the load 120 constituted by the light emitting diode is about 500 mA and the period of the clock signal (clock period) is about 20 to 30 μs, the on period of the switching element T131 is several μs. During this time, the capacitor C132 is discharged, and the voltage VCp at the positive terminal of the capacitor C132 decreases. The amount of decrease in VCp depends on the capacitance Cp of the capacitor C132. That is, as the capacitance Cp increases, the amount of decrease in VCp decreases. When the AC power supply 10 is 100 V and 60 Hz, the peak value of Vdc is 141 V. With respect to this peak value, it is preferable to set the capacitance Cp of the capacitor C132 so that the decrease amount of the positive electrode voltage VCp due to the discharge is several V during the ON period of the switching element T131. The capacitance Cp of the capacitor C132 is set to about 0.3 μf, for example.
On the other hand, when the voltage (load voltage) of the load 120 constituted by a light emitting diode is VLED, when the switching element T131 is turned on, the terminal voltage VL1 of the inductor L131 is expressed by (Expression 2).
VL1 = L1 × (dI2 / dt) = VCp−VLED
(DI2 / dt) = (VCp−VLED) / L1 (Formula 2)
That is, the increase rate (increase gradient) of I2 is proportional to (VCp−VLED) and inversely proportional to L1.

On the other hand, if the time from when the switching element T131 is turned on to when I2 starts to flow and then shut off is ts, Is at the time of I2 shutoff is expressed by (Equation 3).
Is = (dI2 / dt) × ts = (VCp−VLED) / L1 × ts
(Formula 3)
When using (Expression 1) and (Expression 3), the following expression is established.
(R2 / R1) × K × Vdc = (VCp−VLED) / L1 × ts
Since [Vdc≈VCp], the above expression can be expressed as (Expression 4).
ts = (R2 / R1) × K × VCp / (VCp−VLED) × L1
(Formula 4)
Here, in a region where [VCp >> VLED] is satisfied, ts becomes a constant value independent of VCp, and becomes larger than (VCp−VLED) as the value of (VCp−VLED) decreases.
Since the discharge charge amount of the capacitor C132 when the switching element T131 is on (hereinafter referred to as “Cp discharge charge amount”) is [Is × ts / 2], (Equation 5) is satisfied.
Cp discharge charge = Is × (ts / 2)
= (L1 / 2) × [(R2 / R1) × K] ² × VCp ² / (VCp−
VLED) (Formula 5)
VCp is larger than Vdc at the start of discharge and smaller than Vdc at the end of discharge, and the average value during the discharge period is substantially equal to Vdc. Therefore, in (Expression 4) and (Expression 5), [VCp≈Vdc] can be obtained. The amount of Cp discharge charge is proportional to Vdc in a region where [Vdc >> VLED], and becomes larger than the Vdc proportional value as Vdc decreases and approaches VLED.

When the amount of change in Vdc in one cycle (on period + off period) of the switching element T131 is ΔVdc, [ΔVdc> 0] is obtained in the process of increasing Vdc, and [ΔVdc = 0] at the time of Vdc peak. In the decreasing process, [ΔVdc <0]. The capacitor C132 is charged with [ΔVdc × Cp] more charge than the Cp discharge charge amount in the process of increasing Vdc due to the AC input current, and [ΔVdc × Cp] than the Cp discharge charge amount in the process of decreasing Vdc. Only a small amount of charge is charged.
Therefore, (Equation 6) is established in one cycle of the switching element T131. Note that one cycle of the switching element T131 matches the cycle of the clock signal output from the clock signal output circuit 141 (clock cycle).
Cp discharge charge = Cp charge current integrated value−ΔVdc × Cp (Formula 6)
The charge amount [ΔVdc × Cp] acts to advance the phase of the AC input current Iac with respect to the AC voltage Vac, and generates a harmonic current close to the fundamental frequency.
Here, if the value of the capacitance Cp of the capacitor C132 is selected so that [Cp charge current integral value >> (ΔVdc × Cp)], the Cp discharge charge amount is the Cp charge current integral of one cycle of the switching element T131. Almost equal to the value.

As can be understood from FIG. 1, the AC input current Iac is the sum of the charging current of the capacitor C132, the current flowing through the resistor R181, and the current Iref flowing through the resistor R141. Since the current flowing through the resistors R181 and R141 is substantially proportional to Vdc, the AC input current Iac has a magnitude proportional to Vdc. Therefore, the AC input current Iac is substantially in phase with the AC voltage Vac, and the power factor approaches “1”.

Next, the reflux current I3 will be described.
When the switching element T131 is turned off, a counter electromotive force is generated in the inductor L131 in a direction in which the current I2 that was flowing through the inductor L131 when the switching element T131 is turned on is generated, so that the positive terminal voltage of the capacitor C131 increases. Thereby, the negative terminal voltage of the capacitor C131 is raised. When the negative terminal voltage of the capacitor C131 reaches a voltage higher than the positive terminal voltage VCp of the capacitor C132 by the forward voltage of the freewheeling diode D131, the inductor L131, the parallel circuit of the load 120 and the capacitor C131, and the path of the freewheeling diode D131, A reflux current I3 flows. At this time, the back electromotive force generated in the inductor L131 is the sum of the inter-terminal voltage (load voltage) VLED of the capacitor C131 and the forward voltage VD1 of the freewheeling diode D131.
−L1 × (dI3 / dt) = VLED + VD1
− (DI3 / dt) = (VLED + VD1) / L1
The reflux current I3 is proportional to (VLED + VD1) and linearly decreases with a gradient having a magnitude inversely proportional to L1.

[I3 = I2 = Is] is established when the switching element T131 shifts from on to off. As L1 increases, the time until the return current I3 disappears becomes longer. When the time until the return current I3 disappears exceeds the OFF period of the switching element T131, the switching element T131 is turned on while the return current I3 is flowing.
When the value of the return current I3 when the switching element T131 is on is I3rem, the electromagnetic energy newly accumulated in the inductor L131 when the switching element T131 is next off is [L1 × (Is ² −I3rem ² ) / 2]. Become. Also all the electromagnetic energy accumulated in inductor L131 is a ^{[L1 × (Is) 2/} 2], the electromagnetic energy ^{[L1 × (I3rem) 2/} 2] is accumulated constantly inductor L131, is not output. That is, even I3rem flows back circuit, electromagnetic energy ^{[L1 × (I3rem) 2/} 2] is not transmitted to the output side. In the return circuit, power loss occurs due to a drop in the forward voltage of the wiring resistance and the return diode D131. Therefore, the efficiency is lowered due to the non-output I3rem flowing.
On the other hand, the larger the L1, the smaller the current gradient change when changing from I2 to I3, so the loss of the switching element T131 becomes smaller. Therefore, it is desirable to set the inductance L1 as large as possible within a range in which the return current I3 disappears during the OFF period of the switching element T131.

As described above, as a method of changing the magnitude of the DC power supplied to the load 120, a method of changing the resistance value of the resistor R141 or a method of changing the current mirror ratio of T148 and T149 or T146 and T147. Can be used.
If the capacitance Cp of the capacitor C132 is fixed at, for example, 0.3 μF, the power factor can be set within a predetermined range when the DC power is large (can satisfy the PFC standard), but if the DC power is reduced, the power In some cases, the rate cannot be set within a predetermined range (the PFC standard cannot be satisfied). For example, the Cp discharge charge amount decreases and the Cp charge current integral value decreases, and in the above-described (Equation 6), the condition of [Cp charge current integral value >> ΔVdc × Cp] cannot be satisfied.
In such a case, the changeover switch SW131 shown in FIG. 1 is switched to connect the fixed contact e and the normally open contact g. As a result, the capacitor C133 having the capacitance Cp1 is connected in series to the capacitor C132, the combined capacitance (equivalent capacitance of Cp) is reduced, and the condition [Cp charging current integral value >> ΔVdc × Cp] can be satisfied. . For example, if the capacitance Cp1 of the capacitor C133 is set to 0.1 μF, the equivalent capacitance of Cp = 0.075 μF is obtained. When such a capacitor C133 is used, the power factor can be set within a predetermined range even when the resistance value of the resistor R141 is increased.

In addition, in 1st Embodiment 100 shown in FIG. 1, the parallel circuit of the inductor L131, the load 120, and the capacitor | condenser C131 is connected in series. For this reason, even if the position of the parallel circuit of the inductor L131, the load 120, and the capacitor C131 is switched, the obtained function does not change.

Next, the threshold correction circuit 150 will be described.
First, the current I1 is obtained.
The absolute value of AC input current Iac is substantially equal to current I1 flowing through inductor L132. Assume that the voltage at the positive terminal of the capacitor C132 is VCp, and the switching element T131 has shifted from on to off at the time of [t = 0], and the values (initial values) of I1 and VCp at that time are I1 (0) and If VCp (0), I1 is expressed by (Equation 7).

Solving (Equation 7) with Vdc constant, I1 is expressed by (Equation 8).

However,

At time [t = 0], when [Vdc−VCp (0)> 0] and the current I1 starts to increase, the capacitor C132 is charged by I1, and VCp increases.
I1 is a waveform obtained by cutting off a part of a sine wave as represented by (Equation 8). I1 is

At the peak value. At this time, [Vdc−VCp = 0].
Thereafter, VCp increases, but I1 decreases. Then, during the ON period of the next switching element T131, it is inverted from decrease to increase. Then, when the next off period of the switching element T131 starts and I1 starts to increase, one cycle ends.
That is, corresponding to the clock period, one period of the I1 waveform starting from [t = 0] is an upwardly convex waveform composed of a part of a sine wave, and I1 is a convex waveform corresponding to the clock period. It becomes a continuous waveform. The peak value (local maximum value) of the convex waveform has a magnitude proportional to Vdc in the region of [Vdc >> VLED], and becomes a value larger than the Vdc proportional value as (Vdc−VLED) decreases.
The fluctuation range of the convex waveform (difference between the minimum value and the peak value) depends on the ratio of the clock period to the sine wave period. That is, when the period of the sine wave becomes relatively large with respect to the clock period, the fluctuation width of the convex waveform becomes small.

The following points can be considered as the causes of the power factor lowering than “1” and the generation of harmonic current.
(Cause 1): The phase of I1 advances by [ΔVdc × Cp] included in (Expression 6).
(Cause 2): Since I1 is a waveform in which convex waveforms synchronized with the clock cycle are connected, a harmonic current having a clock frequency is generated. The larger the fluctuation range of the convex waveform, the higher the amplitude of the harmonic current.
(Cause 3): Due to variations in the initial values (Vdc−VCp (0)) and I1 (0) included in (Equation 8), the peak value and fluctuation range of I1 are not consistent with a period several times the clock period. It fluctuates regularly and generates a harmonic current with a frequency lower than the clock frequency.
(Cause 4): In the region where (Vdc−VLED) is small, the increasing gradient (increase rate) of I1 becomes gradual, and the ON period of the switching element becomes long. For this reason, even if I1 when the switching element is cut off is proportional to Vdc, the Cp discharge charge amount during the ON period of the switching element is relatively larger than the Vdc proportional value. Therefore, when Vdc increases from the minimum value and the switching element starts to turn on and off, I1 increases stepwise. This triggers a damped oscillation having a period several times the clock period starting from a large amplitude due to (Cause 3).

As a countermeasure for (Cause 2), a method of reducing the clock cycle and relatively increasing the sine wave cycle is effective.
From (Equation 8), the sine wave cycle is [2π × (L2 × Cp) ^1/2 ]. Therefore, in order to increase the sine wave period, L2 or Cp may be increased.
Increasing L2 increases power loss due to internal resistance, product shape and cost. For this reason, the method of increasing L2 is not preferable.
On the other hand, when Cp is increased, the phase advance of I1 due to [ΔVdc × Cp] included in (Expression 6) increases. As the phase advance of I1 increases, the power factor deteriorates.
In order to prevent an increase in the phase advance of I1 due to an increase in Cp, the amount of Cp discharge charge is decreased by [ΔVdc × Cp] when Vdc increases, and the amount of Cp discharge charge is decreased by [ΔVdc when Vdc decreases. XCp] should be increased.
The Cp discharge charge amount is proportional to I14. Therefore, in order to prevent an increase in the phase advance of I1 due to an increase in Cp, when Vdc is increased, I14 is decreased by an amount corresponding to [ΔVdc × Cp], and when Vdc is decreased, I14 is decreased by [ΔVdc × Cp. ] Just increase by a considerable amount.
In this embodiment, by providing a threshold correction circuit 150 that increases or decreases (corrects) I14, while preventing an increase in the phase advance of I1 (deterioration of the power factor), Cp is increased to increase the harmonic current of the clock frequency. The amplitude is reduced.

The threshold correction circuit 150 includes P-type MOSFETs T153 and T154, N-type MOSFETs T151, T152, and T155, a capacitor C151, and diodes D151 and D152.
T151 and T152 and T153 and T154 are arranged to form a current mirror circuit.

The threshold correction circuit 150 operates as follows.
When Vdc increases, a charging current flows through the capacitor C151 through the path of Vdc, capacitor C151, diode D151 and T151. At this time, since the potential of the anode of the diode D151 becomes higher than the ground level, T155 is off.
The current flowing through the capacitor C151 is 90 degrees out of phase with respect to Vdc, and its magnitude is proportional to the differential value of Vdc, and is therefore proportional to [ΔVdc × Cp]. Since the current mirror circuit is constituted by T151 and T152, the drain current I15 of T152 is proportional to the charging current of the capacitor C151, and the magnitude thereof can be adjusted by the current mirror ratio of T151 and T152.
That is, in the increasing process of Vdc, the current I15 proportional to [ΔVdc × Cp] is subtracted from the current I14 proportional to Vdc.
On the other hand, when Vdc decreases, the capacitor C151 pushes down the anode potential of the diode D151 below the ground level, so that T155 is turned on. When T155 is turned on, the discharge current of the capacitor C151 flows through the paths of the capacitors C151, T153, T155, and the diode D152. The discharge current of the capacitor C151 is proportional to [ΔVdc × Cp]. Since the current mirror circuit is configured by T153 and T154, the drain current I16 of T154 is proportional to the discharge current of the capacitor C151, and the magnitude thereof can be adjusted by the current mirror ratio of T153 and T154.
That is, in the process of decreasing Vdc, the current I16 proportional to [ΔVdc × Cp] is added to the current I14 proportional to Vdc.
Accordingly, [Cp discharge charge amount−ΔVdc × Cp] in each clock cycle, that is, the Cp charge amount is proportional to Vdc, and an increase in phase advance due to an increase in Cp can be prevented.

As a countermeasure method for (Cause 3) and (Cause 4), a method in which the switching element is not operated when (Vdc−VLED) after the rise of Vdc is small is effective. This method will be described in a second embodiment (FIG. 3) described later.

The simulation result when the threshold value correction circuit 150 is provided is shown in FIG. The simulation results shown in FIG. 2 are obtained when the values of the circuit elements are set as follows. Vac: AC 100 V, 60 Hz, L1: 50 μH, L2: 600 μH, Cp: 0.6 μF, C1: 500 μF, C151: 1 nF, clock cycle: 20 μs (H level period: 1.6 μs (when Vdc increases), 1.8 μs ( Vdc decrease)), VLED average value: 15 V, LED load current average value: 644 mA.
In FIG. 2, the horizontal axis represents elapsed time t (ms), the first vertical axis represents the voltage (V) of the graph (1), and the second vertical axis represents the graphs (2) and (3). The third axis of the vertical axis represents the current (μA) of graphs (4) to (7).
Graph (1) (solid line with white squares) represents Vdc (V). Graph (2) (solid line with white triangles) represents I1 (mA) when the threshold correction circuit 150 is not provided. Graph (3) (broken line with a black triangle) represents I1 (mA) in the case where the threshold correction circuit 150 is provided. Graph (4) (broken line with a black circle) represents a threshold value (current threshold value) I14 (μA) before correction. A graph (5) (a one-dot chain line with a white square) represents the correction value I15 (μA). Graph (6) (two-dot chain line with a black square) represents the correction value I16 (μA). Graph (7) (solid line with white circles) represents a corrected threshold value (current threshold value) [I14−I15 + I16] (μA).
Although not shown in FIG. 2, the current waveforms shown in graphs (2) to (7) include harmonic current components.

The following can be understood from FIG.
The threshold when the threshold correction circuit 150 is not provided is set by I14 shown in the graph (4). At this time, the phase of I1 (I1 before correction) shown in the graph (2) is advanced with respect to Vdc shown in the graph (1), and the amplitude of the eleventh harmonic current is , Which is 3.32% of the amplitude of the fundamental current. In this case, since the PFC standard cannot be satisfied, a low-pass filter for suppressing harmonic current is required.
On the other hand, when the threshold value correction circuit 150 is provided, the threshold value is the correction value I15 indicated by the graph (5) and the correction value indicated by the graph (6), which is indicated by the graph (4). A graph (7) of a value [I14−I15 + I16] corrected by the value I16 is obtained. At this time, I1 (I1 after correction) shown in the graph (3) suppresses the phase advance with respect to Vdc, and the amplitude of the harmonic current higher than the eleventh harmonic is 50 kHz (clock period: In the case of 20 μs), it was 2.6%, but it was less than this at other frequencies. In this case, since the PFC standard is satisfied, it is not necessary to provide a low-pass filter for suppressing the harmonic current. That is, the harmonic current can be sufficiently suppressed without providing a low-pass filter.

[Second Embodiment]
In the first embodiment, by using the value obtained by correcting the threshold value I14 with the correction values I15 and I16 as the threshold value, the waveform of I1 is modified to bring the power factor closer to “1” (suppress harmonic current). Was used. Other methods can be used as a method of correcting the waveform of I1 to bring the power factor closer to “1”.
FIG. 3 shows a circuit diagram of a second embodiment 200 of the power supply device of the present invention. In the second embodiment 200, a method of changing the threshold value I14 is used.
The second embodiment 200 is different from the first embodiment except that the threshold correction circuit 150 of the first embodiment 100 is not provided in the control circuit 240 and a Zener diode ZD241 is added to the control circuit 240. The configuration is the same as that of the embodiment 100. That is, the configuration of the threshold generation circuit of the control circuit 240 is different from the configuration of the threshold generation circuit of the control circuit 140. In FIG. 3, the components to which the symbols shown in FIG. 1 and symbols other than the numbers in the hundreds are attached are the components of the first embodiment 100 shown in FIG. 1. Is the same.

Only the configuration (configuration of the threshold generation circuit) that is different from the first embodiment 100 will be described below.
The threshold value generation circuit of the present embodiment 200 includes T246 to T249, a resistor R241, and a Zener diode ZD241.
If the resistance value of the resistor R241 is R241a and the Zener voltage of the Zener diode ZD241 is VZD241, Iref is expressed by the following equation.
Iref = (Vdc−VZD241) / R241a
Note that I14 is proportional to Iref.
In the case of [Vdc <VZD241], since Iref does not flow, I14 becomes zero. At this time, if the switching element T231 is configured to be turned off, I1 does not flow, so that the positive terminal voltages VCp and Vdc of the capacitor 232 do not decrease below the VZD241.
On the other hand, when [Vdc> VZD241], Iref flows. At this time, Iref is proportional to (Vdc−VZD241).

Now, a case where the amplitude of the AC voltage Vac is set to 141 V and the VZD 241 is set to 47 V (= 141 V / 3) will be described.
In this case, the change range of VCp in a half cycle of the AC voltage Vac is 47 V to 141 V, and the sum of absolute values of [ΔVdc × Cp] is [(141 V−47 V) × Cp × 2]. On the other hand, when the Zener diode ZD241 is not provided, the sum of absolute values of [ΔVdc × Cp] is [141V × Cp × 2].
That is, in the case where the Zener diode ZD241 is provided, the charge / discharge charge amount of Cp due to the change in Vdc is 2/3 compared to the case where the Zener diode ZD241 is not provided, and the phase advance due to Cp is increased. Decrease to 2/3.
Therefore, the phase advance in the case where the Zener diode ZD241 is provided (second embodiment) and Cp is set to 0.6 μf, the Zener diode ZD241 is not provided and Cp is set to 0.4 μf. It is the same as the phase lead when In other words, in this embodiment, Cp can be increased while suppressing an increase in phase advance due to an increase in Cp.

In the case where the Zener diode ZD241 is not provided, the integral value of Iref and the half cycle of Vdc is proportional to the area of the half cycle of Vdc.
On the other hand, in the case where the Zener diode ZD241 is provided, the integral value of the half cycle of Vdc of Iref flowing in the region of [Vdc> VZD241] is a portion where Vdc is VZD241 or less from the area of the half cycle of Vdc. Is proportional to the area obtained by subtracting (subtracting) the area. When this area is calculated for the case where VZD 241 is set to 47 V, it becomes 53.25% of the area of the half cycle of Vdc. For this reason, when the resistance value of the resistor R241 is the same as the case where the Zener diode ZD241 is not provided (for example, the same as the resistance value of the resistor R141 of the first embodiment 100), the integration of Vref half cycle of Iref is performed. The value is smaller than when the Zener diode ZD241 is not provided.
In order to make the integral value of Vref half cycle of Iref equal to the case where the Zener diode ZD241 is not provided, the resistance value R241a may be reduced. For example, when the resistance value of the resistor R141 of the first embodiment 100 is R141a, the resistance value R241a may be set to [R241a = R141a × 0.5325].
Even if the resistance value R241a is changed (decreased), the phase advance of I1 does not increase even though it decreases.
If resistance value R241a is further reduced, I1 increases and the amplitude of the fundamental current increases. In this case, the amplitude ratio of the harmonic current to the amplitude of the fundamental wave current becomes small.
Further, the correlation between Iref and (Vdc−VZD241) can be set to a correlation different from the proportionality. For example, a circuit in which a resistor and a Zener diode are connected in series is connected in parallel with the resistor R241.

The simulation was performed by setting the values of the circuit elements of the second embodiment 200 as follows.
Vac: AC 100 V, 60 Hz, L1: 50 μH, L2: 600 μH, Cp: 0.6 μF, C1: 500 μF, clock cycle: 20 μs (H level period: 2 μs (when Vdc increases), 2 μs (when Vdc decreases)), VZD241: 48V, VLED average value: 15.4V, LED load current average value: 729 mA.
As a result of the simulation, the power factor was 0.98 or more. The amplitude of the 11th to 39th harmonic currents is 2% or less of the amplitude of the fundamental wave current, which satisfies the PFC standard. The amplitude of the harmonic current above the 40th harmonic was 3.7% when the frequency was 8.3 kHz and 2.6% when the frequency was 50 kHz, but it was less than these at other frequencies. It was. In this case, since the PFC standard is satisfied, it is not necessary to provide a low-pass filter for suppressing the harmonic current.
That is, by providing the Zener diode ZD241 in the threshold generation circuit, it is possible to obtain a phase advance suppression effect equivalent to that of the threshold correction circuit 150 of the first embodiment 100.
Note that the power factor correction method of the second embodiment can also be used in combination with the threshold correction circuit 150 of the first embodiment. In this case, for example, in FIG. 1, a Zener diode is added between the resistors R141 and T149 to decrease the resistance value of the resistor R141.

[Third Embodiment]
FIG. 4 shows a circuit diagram of a third embodiment 300 of the power supply device of the present invention.
In the third embodiment 300, the inductor L131 of the first embodiment 100 is removed, and the inductor L331 is disposed between the inductor L332 and the capacitor C332. The inductor L331 has the same inductance L1 as the inductance L1 of the inductor L131 of the first embodiment 100. The inductance L2 of the inductor L332 is set to be larger than the inductance L1 of the inductor L331 as in the first embodiment 100.
In FIG. 4, the constituent elements to which the symbols shown in FIG. 1 and symbols other than the numbers in the hundreds are the same are the constituent elements of the first embodiment shown in FIG. The same thing.
FIG. 4 does not show the threshold correction circuit of the first embodiment 100 or the threshold generation circuit of the second embodiment for correcting the waveform of I1 to bring the power factor close to “1”. Of course, at least one of the threshold value correction circuit and the threshold value generation circuit may be provided. This is the same in other embodiments described below.

The third embodiment 300 operates as follows.
When the switching element T331 is off, the charging current I1 of the capacitor C332 flows through the full-wave rectifier circuit 310, the inductor L332, the inductor L331, the capacitor C332, the fixed contact e of the changeover switch SW331, the normally closed contact f, and the ground path. .
When the switching element T331 is turned on, the capacitor C332, the inductor L331, the parallel circuit of the load 320 and the capacitor C331, the switching element T331, the current detection resistor R331, the ground and the switch SW331 fixed contact e and the normally closed contact f, A discharge current I2 of the capacitor C332 flows.
At this time, the current I4 (not shown) also flows through the full-wave rectifier circuit 310, the inductor L332, the parallel circuit of the load 320 and the capacitor C331, the switching element T331, the current detection resistor R331, and the ground path. However, when the condition [inductance L2 of inductor L332 >> inductance L1 of inductor L331] is satisfied, [I2 >> I4] is satisfied.

Also in the first embodiment 100 shown in FIG. 1, when the switching element T131 is turned on, the full-wave rectifier circuit 110, the inductor L132, the inductor L131, the parallel circuit of the load 120 and the capacitor C131, the switching element T131, and the current detection resistor A current I5 (not shown) flows through the path of R131 and the ground. Also in this case, when the condition [inductance L2 of inductor L132 >> inductance L1 of inductor 131] is satisfied, [I2 >> I5] is satisfied.
Since the current I5 flows through the inductors L132 and L131, and the current I4 flows only through the inductor L232, [I4> I5] is satisfied. However, since I4 and I5 are smaller than I2, the circuit shown in FIG. 1 and the circuit shown in FIG. 4 can be regarded as equivalent circuits.

The difference between the first embodiment 100 and the third embodiment 300 is that the amount of decrease in potential at the point r or t when the switching element is turned on is different.
In the first embodiment 100, when the switching element T131 is turned on, the potential at the point r becomes the potential of the positive terminal of the capacitor C132, and therefore the amount of decrease is smaller than that in the third embodiment. Thereby, the fluctuation | variation of the electric current which flows through the inductor L132 decreases.
However, the potential at the point r when the switching element T131 is on is likely to fluctuate. For this reason, the periodicity of the current flowing through the inductor L132 is disturbed, and fluctuations with a period longer than the clock period are likely to occur.
On the other hand, in the third embodiment 300, the potential at the point t when the switching element T331 is on is the load voltage VLED, and there is little fluctuation. Thereby, it is excellent in the periodicity of the electric current which flows through the inductor L332. On the other hand, since the amount of decrease in the potential at the point t when the switching element T331 is on is large, the amplitude of the harmonic current of the clock frequency becomes large.

[Fourth Embodiment]
In the first to third embodiments, the switching element is disposed between the other end of the load and the negative end of the full-wave rectifier circuit. However, the switching element is disposed between one end of the load and the positive end of the full-wave rectifier circuit. It can also be arranged.
FIG. 5 shows a circuit diagram of a fourth embodiment 400 of the power supply device of the present invention. In the fourth embodiment 400, a switching element T431 and a current detection resistor R431 are arranged in series between one end 431 of the load 420 and the positive end c of the full-wave rectifier circuit 410.
The control circuit 440 includes P-type MOSFETs T441 to T443, T448 and T449, N-type MOSFETs T444 to T447, resistors R441 to R444, constant current source IDC441, clock signal generation circuit 441, Schmitt trigger ST451, and D flip-flop. The circuit includes a DFF 441, an AND circuit AND441, and a drive circuit 442.
The power supply circuit 430, the control circuit 440, the power supply circuit 480, and the drive signal output circuit 490 control the switching element T431 disposed between one end 431 of the load 420 and the positive end c of the full-wave rectifier circuit 410. It is configured to be able to.

The operation of the fourth embodiment 400 is similar to the operation of the first embodiment 100.
When the switching element T431 is off, the charging current of the capacitor C432 flows through the full-wave rectifier circuit 410, the capacitor C432, the inductor L432, and the ground path.
When the switching element T431 is turned on, the discharge current of the capacitor C432 flows through the capacitor C432, the current detection resistor R431, the switching element T431, the parallel circuit of the load 420 and the capacitor C431, and the path of the inductor L431.
When the voltage drop (current flowing through T431) of the current detection resistor R431 becomes larger than the threshold value, the switching element T431 is turned off. At this time, the return current flows through the path of the inductor L431, the return diode D431, and the parallel circuit of the load 420 and the capacitor C431 due to the electromagnetic energy accumulated in the inductor L431.
The inductance L1 of the inductor L431 is set to such a magnitude that the return current disappears during the OFF period of the switching element T431. Further, the inductance L2 of the inductor L432 is set to such a magnitude that the charging current of the capacitor C432 continues to flow during the OFF period of the switching element T431.

In the fourth embodiment 400, since the switching element T431 is disposed between one end (positive electrode end) 431 of the load 420 and the positive electrode end c of the full-wave rectifier circuit 410, the wiring downstream of the switching element T431 is grounded. Thus, when an overcurrent flows, the switching element T431 and the downstream wiring can be protected.
In the fourth embodiment 400, the positions of the inductor L431, the parallel circuit of the load 420 and the capacitor C431 may be switched.

[Fifth Embodiment]
The efficiency of the first to fourth embodiments is determined by the loss of the diode used in the full-wave rectifier circuit, the switching loss of the switching element, the loss of the current detection resistor, the loss of the freewheeling diode, and the like. Among these losses, the loss of the freewheeling diode and the diode of the full wave rectifier circuit is large. In particular, the loss of the return diode is large. This is because the current (I1) flowing through the diode of the full-wave rectifier circuit is smaller than the return current (I3) flowing through the freewheeling diode. For example, when the voltage between the terminals of the load is 20V and the effective value of the AC voltage is 100V, [I1 / I3≈20V / 100V = 1/5]. I1 flows through two diodes connected in series, and I3 flows through one freewheeling diode. For this reason, the loss of the diode of the full-wave rectifier circuit is about (1 / 2.5) of the loss of the freewheeling diode.
In the fifth embodiment, in order to reduce the loss when the return current flows, the return diode used in the first to fourth embodiments is replaced with a MOSFET parasitic diode.

A circuit diagram of the fifth embodiment 500 is shown in FIG.
In the fifth embodiment 500, the freewheeling diode D131 of the first embodiment 100 is replaced with an FT532 that is an N-type MOSFET, and a freewheeling current is passed through a parasitic diode of the FT532. “FT532” is a simple symbol for distinguishing an FET used as a freewheeling diode from other FETs. In addition, a voltage detection circuit 560 and a drive circuit 570 are added.
The fifth embodiment 500 has the same configuration as the first embodiment 100 except for the FT 532, the voltage detection circuit 560, the drive circuit 570, and the power supply circuit 580.
In FIG. 6, the constituent elements to which the symbols shown in FIG. 1 and the symbols other than the numbers in the hundreds are the same are the constituent elements of the first embodiment 100 shown in FIG. Is the same. FIG. 6 shows only the main part of the fifth embodiment 500.
Below, FT532 of this embodiment is mainly demonstrated.

The power supply circuit 580 includes an N-type MOSFET T581, a resistor R581, a Zener diode ZD581, and a capacitor C581, which are disposed between one end 531 and the other end 532 of the load 520. The power supply circuit 580 supplies power to the voltage detection circuit 560 that detects the voltage between the drain and source of the FT 532 (the voltage between the anode and cathode of the parasitic diode) and the drive circuit 570 that drives the FT 532.

The voltage detection circuit 560 includes P-type MOSFETs T561, T563, and T566, N-type MOSFETs T562, T564, T565, and T567, resistors R561 to R564, and a DC power supply V560.
Since the voltage between the terminals of the resistor R561 becomes a constant voltage, the current flowing through the resistor R561 becomes a constant current.
T567 and DC power supply V560 are drain voltage inputs for preventing a large voltage generated between the drain and source of FT532 from being input to the source of T564 via resistors R564 and R563 when switching element T531 is turned on. A limiting circuit is configured.
FETs having the same characteristics are used as T561 and T563, and FETs having the same characteristics are used as T562 and T564. In addition, resistors having the same resistance value are used as the resistors R562 and R563.

The drive circuit 570 includes P-type MOSFETs T571, T573, and T574, N-type MOSFETs T572, T575, and T576, resistors R571 to R576, a diode D561, a Zener diode ZD561, a Schmitt trigger ST571, and an astable multivibrator 1Shot571. Has been.
The voltage at point s in FIG. 1 is input to the trigger terminal Trig of 1 Shot 571 via ST571. 1Shot 571 is reset when the voltage at point s input to the reset terminal R is at L level.

The operation of this embodiment will be described.
A case where the resistance value of the resistor R564 is 0Ω will be described.
When the switching element T531 is turned off, a return current flows due to electromagnetic energy accumulated in the inductor L531.
When the parasitic diode of FT532 is forward-biased by the return current, the drain voltage of FT532 becomes lower than the source voltage, and T564 can pass a current larger than the drain current of T562. Thereby, the voltage of the connection point of T563 and T564 falls. Then, the voltage at the connection point w between the source of T565 and the source of T566 decreases, and T576 is turned off. At the same time, T572 is turned off and T571 is turned off.

On the other hand, when the switching element T531 shifts from on to off, the s-point voltage changes from the L level to the H level. When this change is input to the trigger terminal Trig of 1 Shot 571 via ST571, the non-inverted output + Q becomes H level for a certain time. The “certain time” is set to a time shorter than the time from when the reflux current starts to flow until it disappears. For example, when the clock period is about 30 μs, it is preferably set to about 3 μs.
Thereby, after the switching element T531 is turned off, T575 is turned on for about 3 μs, and the resistor R575 is grounded. At this time, the source voltage of the FT532 becomes higher than the positive terminal voltage VCp of the capacitor C532. In this state, the resistance values of the resistors R574 and R575 can be set such that the gate voltage of T574, that is, the connection voltage of the resistors R574 and R575 is lower than the source voltage of the FT532 at any value of Vdc. .
When the resistance values of the resistor R574 and the resistor 575 are set in this manner, the switching element T531 is turned off, and then the T574 is turned on for about 3 μs. Thereby, T573 can be turned on during this period, the gate of FT532 is charged, and FT532 is turned on.
A certain time during which the non-inverted output + Q of 1 Shot 571 becomes H level corresponds to the “predetermined period” of the present invention.

When the return current monotonously decreases, the forward voltage of the parasitic diode of FT532 decreases, and the difference approaches zero in the state of [drain voltage <source voltage]. As a result, the voltage at the point w increases and T576 is turned on. At this point, since T575 is already off, even if T571 is off, the source-gate voltage of T573 becomes zero by the resistor R576. Thereby, since T573 maintains an off state, FT532 is turned off.
When the resistance R564 is 0Ω, the w-point voltage starts to rise when the return current reaches the zero cross. Here, since there is an FT532 OFF delay, the FT532 is substantially turned OFF at a timing when the return current exceeds the zero cross and starts to flow in the reverse direction. When turned off at this timing, back electromotive force is generated in the inductor L531, the inductance of the inductor L531 and the capacitance between the drain and source of the FT532 resonate, and the drain-source voltage of the FT532 is hunted. To avoid this, the FT 532 may be cut off before the return current reaches the zero cross. In order to realize this, a resistor R564 is provided. As the resistance value of the resistor R564 is increased, the cutoff timing is advanced.

In summary, when the switching element T531 is turned off and the return current starts to flow, T575 is turned on for about 3 μs. At the same time, the w point voltage, which is the output of the voltage detection circuit 560 that detects the magnitude of the source-drain voltage of the FT 532, decreases, and T576 and T572 are turned off. At this time, since T575 is on, T573 is on. As a result, a positive voltage is applied to the gate of FT532, the gate is charged, and FT532 is turned on.
When about 3 μs elapses, T575 is turned off, so that T573 is turned off and voltage application to the gate of FT532 is stopped. Even in this state, there is no path for discharging the charge accumulated in the gate of FT532. For this reason, the gate of FT532 is held at a voltage higher than that of the source, and FT532 is kept on.
When the return current approaches the zero cross, the output (voltage at the point w) of the voltage detection circuit 560 increases and T576 is turned on. As a result, the charge accumulated in the gate of FT532 is discharged, so that FT532 is turned off. That is, the FT 532 is turned on immediately after the switching element T531 is turned off, is kept on until just before the zero cross, and is turned off before the zero cross.
Thereby, the switching element T531 is turned off, and the loss when the return current flows through the FT532 by the inductor L531 can be reduced.

FIG. 7 shows the waveforms of the respective parts when the fifth embodiment 500 is simulated.
The simulation conditions are as follows.
AC voltage Vac: 100 V, 60 Hz, inductor L531 inductance L1: 70 μH, inductor L532 inductance L2: 600 μH, capacitor C531 capacitance C1: 500 μF, capacitor C532 capacitance Cp: 0.3 μF, load 520: LED stack (6 pieces) (Series × 6 parallel), resistance value R1 of current detection resistor R531, 0.08Ω, resistance value of resistor R561: 300kΩ, resistance value of resistor R562 and resistance value of resistor R563: 2kΩ, resistance value of resistor R564: 500Ω, constant Current source IDC 141 current 11: 40 μA, clock period: 30 μs, 1 Shot 571 timer time: 3 μs, Zener diode ZD581 Zener voltage: 18 V, Capacitor C581 capacitance C2: 1 μF.

In FIG. 7, the horizontal axis represents the elapsed time t (ms), the first vertical axis represents the voltage (V) of the graphs (1) to (3), and the second vertical axis represents the graph (4). (6) represents the voltage (V), and the third axis of the vertical axis represents the current (A) in the graph (7).
Further, the graph (1) (the one-dot chain line with a white circle) represents the positive voltage (V: first axis) of the capacitor C531. Graph (2) (two-dot broken line with black circles) represents the negative voltage (V: first axis) of capacitor C531. Graph (3) (broken line with a white triangle) represents the positive terminal voltage VCp (V: first axis) of the capacitor C532. A graph (4) (a dashed-dotted line with a white square) represents a source-drain voltage (V: second axis) of T574. A graph (5) (a dashed-dotted line with a black square) represents a gate-source voltage of FT532 (V: second axis). Graph (6) (broken line with a black triangle) represents [w point−source voltage of FT532] (V: second axis). Graph (7) (solid line) represents current IL1 (A: third axis) flowing through inductor L531.

The following can be understood from the waveform of FIG.
At the elapsed time [30.86 ms], the switching element T531 is turned on, and the current flowing through the inductor L531 increases linearly. At this time, most of the current IL1 of the inductor L131 is the discharge current I2 of the capacitor C532. Due to the discharge of the capacitor C532, the positive terminal voltage VCp of the capacitor C532 decreases.
When IL1 reaches 5A, the switching element T531 is turned off. As a result, the negative terminal voltage of the capacitor C531 that has dropped to the ground level when the switching element T531 is turned on rises rapidly due to the back electromotive force of the inductor L531, and exceeds the positive terminal voltage VCp of the capacitor C532. Thereby, the return current I3 flows through the path of the inductor L531, the parallel circuit of the load 520 and the capacitor C531, and the parasitic diode of the FT532. At this time, the negative terminal voltage of the capacitor C531 is clamped to a voltage slightly exceeding the positive terminal voltage VCp of the capacitor C532.
1 Shot 571 turns on T574, and the source-drain voltage of T574 becomes zero for 3 μs after the switching element T531 is turned off. During this time, a forward current flows through the parasitic diode of the FT 532, and [w point−the voltage between the sources of the FT 532] decreases to zero. As a result, T576 is turned off and T573 is turned on, so that the gate and the source of FT532 are charged and FT532 is turned on.
When the return current IL1 due to the inductor L531 monotonously decreases and approaches the zero cross, [w point−the voltage between the sources of the FT532] increases. Thereby, T576 is turned on, the charge of the gate of FT532 is discharged, and FT532 is turned off.
After the current IL1 of the inductor L531 becomes zero, the negative terminal voltage of the capacitor C531 is lower than the positive terminal voltage VCp of the capacitor C532, and one cycle is completed.

[Sixth Embodiment]
In the first embodiment 100 shown in FIG. 1, the function does not change even if the position of the parallel circuit of the inductor L131, the load 120, and the capacitor C131 is changed. However, in this case, the configuration shown in FIG. 6 cannot be used to replace the free wheel diode with the parasitic diode of the MOSFET. In the sixth embodiment, another configuration for replacing the freewheeling diode with a parasitic diode of the MOSFET is used. FIG. 8 shows a circuit diagram of the sixth embodiment 600.
The sixth embodiment 600 has the same configuration as that of the first embodiment 100 except for the FT 632, the voltage detection circuit 660, the drive circuit 670, and the power supply circuit 680.
In FIG. 8, the components to which the symbols shown in FIG. 1 and symbols other than the numbers in the hundreds are the same are the components of the first embodiment 100 shown in FIG. 1. Is the same. Further, FIG. 8 shows only the main part of the sixth embodiment 600.

In the present embodiment 600, the inductor L631 is arranged on the negative electrode side from the parallel circuit of the load 620 and the capacitor C631. For this reason, a P-type MOSFET is used as the FT 632 to be replaced with the freewheeling diode. The circuit for controlling the FT 632 of the sixth embodiment 600 is different in configuration from the circuit for controlling the FT 532 of the fifth embodiment 500, but the concept is the same.

The power supply circuit 680 includes a P-type MOSFET T681, a resistor R681, a Zener diode ZD681, and a capacitor C681. The circuit that controls the FT 632 operates within a range lower than the Zener voltage of the Zener diode ZD681 with reference to the positive terminal voltage VCp of the capacitor C632.
The voltage detection circuit 660 includes P-type MOSFETs T661, T663, T666, and T667, N-type MOSFETs T662, T664, and T665, resistors R661 to R664, and a DC power supply V660.
The drive circuit 670 includes P-type MOSFETs T671 to T673, N-type MOSFETs T674 to T676, resistors R671 to R676, a Zener diode ZD671, a Schmitt trigger ST671, and an astable multivibrator 1Shot671.

The operation of the present embodiment 600 will be described.
When the potential at point s in FIG. 1 rises and an H level signal is input to the trigger terminal Trig of 1 Shot 671 via ST671, the non-inverted output + Q becomes H level for about 3 μs. Thereby, T676 and T672 are turned on.
At the same time, the parasitic diode of the FT 632 is forward-biased, and a return current by the inductor L631 flows. Then, the drain voltage of the FT 632 becomes larger than the source voltage by the forward voltage drop of the parasitic diode. As a result, the potential at the point x increases, T671 and T673 are turned off, and T674 is turned off. Since T672 is turned on, T675 is turned on, the gate voltage of FT632 is lower than the source voltage, and FT632 is turned on.
When 3 μs elapses after the switching element T631 is turned off, T672 is turned off. At this time, since the gate charge of the FT 632 remains, the FT 632 is kept on.
When the reflux current monotonously decreases and approaches the zero cross, the potential at the point x decreases and T673 is turned on. At the same time, since T671 and T674 are turned on, T675 remains off and FT632 is turned off. That is, the FT 632 is turned on immediately after the switching element T631 is turned off, and turned off before the return current reaches the zero cross. Thereby, the power loss of FT632 by return current can be reduced.

[Seventh Embodiment]
FIG. 9 shows a circuit diagram of a seventh embodiment 700 using another configuration for replacing the free wheel diode with a MOSFET parasitic diode.
The seventh embodiment 700 has the same configuration as that of the first embodiment 100 except for the FT 732, the voltage detection circuit 760, and the drive circuit 770.
In FIG. 9, the components to which the symbols shown in FIG. 1 and symbols other than the numbers in the hundreds are the same are the components of the first embodiment 100 shown in FIG. 1. Is the same. FIG. 9 shows only the main part of the seventh embodiment 700.
In this embodiment, the freewheeling diode is replaced with a parasitic diode of FT732 which is an N-type MOSFET.

The voltage detection circuit 760 includes P-type MOSFETs T761 and T763, N-type MOSFETs T762 and T764, resistors R761 to R764, diodes D761 and D762, and a Zener diode ZD761.
A first series circuit in which a diode D762, resistors R763 and T763, resistors R764 and T764 are arranged in series is arranged between the drain of the FT732 and the ground, and a current corresponding to the magnitude of the voltage between the drain of the FT732 and the ground It is configured to flow.
When a second series circuit in which a diode D761, resistors R761, T761, resistors R762 and T762 are arranged in series is arranged between the source of FT732 and the ground, and the parasitic diode of FT732 is forward-biased, the source of FT732 -It is comprised so that the electric current according to the voltage difference between drains may flow.
FETs having the same characteristics are used as T761 and T763, and FETs having the same characteristics are used as T762 and T764.
When the current that can flow through the second series circuit is equal to the current in the first series circuit, the voltage drop generated in the resistor R762 is the voltage drop generated in the resistor R764. It is set to be smaller. In this case, the resistance value of the resistor R762 is smaller than the resistance value of the resistor R764.

When the resistance value of the resistor R761 is equal to the resistance value of the resistor R763, if the drain-source voltage of the FT732 is zero, the current flowing through the first series circuit is equal to the current flowing through the second series circuit. On the other hand, when the source voltage of FT732 becomes larger than the drain voltage, a larger current can flow in the second series circuit than in the first series circuit. At this time, the source-drain voltage of T762 increases due to the principle of active load. As a result, the voltage at the connection point z between T762 and the resistor R762 increases, and the output of the Schmitt trigger ST772 becomes H level. At this time, a current having the same magnitude as the current flowing through the first series circuit flows through the second series circuit. Thereby, the expansion of the drain-source voltage of T762 is limited by the voltage drop generated in the resistor R762. The limit width of the increase of the drain-source voltage at T762 becomes smaller as the difference between the resistance value of the resistor R762 and the resistance value of the resistor R764 is smaller.
When the resistance value of the resistor R761 is set to a value slightly larger than the resistance value of the resistor R763, when the source voltage of the FT732 approaches the drain voltage in a range larger than the drain voltage, before the drain-source voltage becomes zero, The current of the series circuit 2 becomes smaller than that of the first series circuit, and T762 is saturated. As a result, the drain-source voltage at T761 increases, the voltage at point z decreases, and the output at ST772 goes to L level.

When the switching element T731 is turned on, the source voltage of the FT732 decreases to the ground voltage. As a result, no current flows through the second series circuit, and the voltage drop across the resistor R762 becomes zero.
When T762 is saturated, the drain voltage of T761 becomes equal to the drain voltage of T762. Thereby, the source-drain voltage of T761 is expanded to the positive terminal voltage VCp of the capacitor C732. The maximum value approaches the peak value of the AC voltage.
In order to prevent this, a Zener diode ZD761 is arranged between the drain of FT732 and the drain of T761 to limit the expansion of the source-drain voltage of T761.

The drive circuit 770 includes N-type MOSFETs T771 and T773 to T776, P-type MOSFETs T772, resistors R771 and R772, Schmitt triggers ST771 and ST772, an astable multivibrator 1 Shot 771, an AND circuit AND771, and a DC power supply V770. ing.
T771 and T772 are arranged to constitute a push-pull circuit.

The operation of the embodiment 700 will be described.
When the switching element T731 is turned on, the source of T732 is grounded, and the potential at the connection point z between the resistors R762 and T762 is lowered. As a result, the output of ST772 becomes L level and the output of AND771 becomes L. As a result, T776 is turned off, and a constant current I8 flows through T775. Further, a current equal to the current I8 flows through the resistors R771, T773, and T774, T771 is turned off, T772 is turned on, and FT732 is turned off.
When the switching element T731 is turned off, the potential at the point s rises, and the output + Q of 1 Shot 771 becomes the H level for a certain period.
When the switching element T731 is turned off, the return current I3 flows. Thereby, the parasitic diode of FT732 is forward-biased, and the source voltage of FT732 becomes larger than the drain voltage.
When the potential at point z rises, the output of ST772 becomes H level. As a result, the output of AND771 becomes H level and T776 is turned on. When T776 is turned on, the current I8 flowing through T775 is cut off, and the current flowing through the resistor R771 becomes zero. Thereby, T771 is turned on, T772 is turned off, and FT732 is turned on.
When the reflux current I3 decreases to zero, the potential at the z point decreases. As a result, the output of ST772 becomes L level, T776 is turned off, and FT732 is turned off.
Since there is a delay time until the potential at the z point decreases and the FT 732 is turned off, the potential at the z point is preferably lowered before the source-drain voltage of the FT 732 becomes zero. For this purpose, the resistance value of the resistor R761 needs to be set slightly larger than the resistance value of the resistor R763.

The voltage of the DC power supply V770 is set to about 6V. In this case, since the source voltage of T773 whose gate is connected to the DC power supply V770 does not exceed 6V, the breakdown voltage of T774 may be about 10V.
Among the FETs used in the circuit for controlling the FET used as the free-wheeling diode, T761 and T762 that may increase the drain-source voltage can be protected by the resistor R762 and the Zener diode ZD761. It can be protected with T773. For this reason, the only FETs that require a large breakdown voltage are FT732 and T773.

In the seventh embodiment 700, each circuit is configured based on GND. And the breakdown voltage of FET used can be made into a voltage lower than an alternating voltage except FT732 and T773. As a result, these circuits (except for FT732 and T773) can be incorporated in the same IC chip as the circuit for controlling on / off of the switching element T731.

The present invention is not limited to the configuration described in the detailed description, and various modifications, additions, and deletions can be made without departing from the gist of the present invention.
Although the case where DC power is supplied to a load having a light emitting diode has been described, the power supply apparatus of the present invention can supply DC power to various loads other than the light emitting diode.
It is preferable to use the threshold value correction circuit shown in FIG. 1 or the threshold value generation circuit shown in FIG. 3 to correct the waveform of I1 and bring the power factor close to “1”. The correction circuit and the threshold generation circuit can be omitted. Even in this case, the power factor can be improved and the efficiency can be improved.
Each structure demonstrated by embodiment can also be used independently, and can also be used combining the plurality selected suitably.
The values (for example, inductance, capacitance, resistance value) of elements constituting each circuit can be appropriately set according to the type of load.
As the switching element for supplying power to the load, an FET is preferably used, but an element other than the FET can also be used.

DESCRIPTION OF SYMBOLS 10 ... AC power source, 100, 200, 300, 400, 500, 600, 700, 800 ... Power supply device, 110, 210, 310, 410, 810 ... Full wave rectifier circuit, 120, 220, 320, 420, 520, 620 , 720, 820... Load, 121, 221, 321, 421, 521, 621, 721, 821... Light emitting diode (LED), 130, 230, 330, 430, 830. , 840 ... control circuit, 141, 241 ... clock signal generation circuit, 142, 242, 442, 560, 660, 760 ... drive circuit, 150 ... threshold correction circuit, 180, 280, 380, 480, 580, 680 ... power supply circuit , 190, 290, 390, 490 ... drive signal output circuit, 560, 660, 760 ... voltage detection Road, 900 ... low-pass filter.

Claims (8)

1. A power supply apparatus comprising: a rectifier circuit that generates a DC voltage obtained by rectifying an AC voltage between a positive electrode end and a negative electrode end; a power supply circuit provided between the rectifier circuit and a load; and a control circuit that controls the power supply circuit Because
   Between the positive electrode end and the negative electrode end of the rectifier circuit, a second inductor and a second capacitor are arranged in series,
   Between the terminals of the second capacitor, a first inductor, a parallel circuit of the first capacitor and the load, and a switching element are arranged in series,
   A diode is arranged in parallel in a series circuit of the first inductor and the parallel circuit of the first capacitor and the load,
   The inductance of the second inductor is set to a value larger than the inductance of the first inductor,
   The control circuit is configured to turn on the switching element at a set cycle, and to turn off the switching element when a current flowing through the switching element becomes larger than a threshold linked to the magnitude of the DC voltage,
   When the switching element is off, the second capacitor is charged. When the switching element is on, the second capacitor is a parallel circuit of the first inductor, the first capacitor, and the DC load. When the switching element is discharged and when the switching element is turned off, electromagnetic energy accumulated in the first inductor when the switching element is turned on is generated between the first capacitor and the DC load. A power supply device configured to be converted into a return current flowing through a path including a parallel circuit and the diode.
2. A power supply apparatus comprising: a rectifier circuit that generates a DC voltage obtained by rectifying an AC voltage between a positive electrode end and a negative electrode end; a power supply circuit provided between the rectifier circuit and a load; and a control circuit that controls the power supply circuit Because
   Between the positive terminal and the negative terminal of the rectifier circuit, a second inductor, a first inductor, and a second capacitor are arranged in series,
   In the series circuit of the first inductor and the second capacitor, a parallel circuit of the first capacitor and the load and a switching element are arranged in series,
   A diode is arranged in parallel in a series circuit of the first inductor and the parallel circuit of the first capacitor and the load,
   The inductance of the second inductor is set to a value larger than the inductance of the first inductor,
   The control circuit is configured to turn on the switching element at a set cycle, and to turn off the switching element when a current flowing through the switching element becomes larger than a threshold linked to the magnitude of the DC voltage,
   When the switching element is off, the second capacitor is charged. When the switching element is on, the second capacitor is a parallel circuit of the first inductor, the first capacitor, and the DC load. When the switching element is discharged and when the switching element is turned off, electromagnetic energy accumulated in the first inductor when the switching element is turned on is generated between the first capacitor and the DC load. A power supply device configured to be converted into a return current flowing through a path including a parallel circuit and the diode.
3. The power supply device according to claim 1,
   The control device executes processing for each clock cycle, and when the DC voltage is in an increasing process, the control device changes the charge amount of the second capacitor according to the increase in the DC voltage during the clock cycle from the threshold value. A value obtained by subtracting a corresponding value is set as the threshold value, and when the DC voltage is in a decreasing process, the threshold value corresponds to the discharge charge amount of the second capacitor due to the decrease in the DC voltage during the clock period. A power supply device configured to use a value obtained by adding values as the threshold value.
4. The power supply device according to claim 1 or 3,
   The threshold is configured to be linked to a value obtained by subtracting a set value from the magnitude of the DC voltage,
   The control device is configured to turn off the switching element when the magnitude of the DC voltage is smaller than the set value.
5. The power supply device according to any one of claims 1 to 4,
   The power supply circuit includes a third capacitor that can be arranged in series with the second capacitor.
6. The power supply device according to any one of claims 1 to 5,
   As the diode, a parasitic diode of MOSFET is used,
   The control circuit turns on the MOSFET when the cathode voltage of the parasitic diode becomes smaller than a value obtained by subtracting the first set value from the anode voltage within a set period after the switching element is turned off. The power supply device is configured to turn off the MOSFET when a cathode voltage of the parasitic diode becomes larger than a value obtained by subtracting the first set value from an anode voltage after a set period has elapsed. .
7. The power supply device according to claim 6,
   A voltage detection circuit for detecting a voltage between the anode and cathode of the parasitic diode;
   The voltage detection circuit includes a first source follower circuit and a second source follower circuit configured by FETs, and a source resistance of the first source follower circuit is connected to an anode of the parasitic diode, The source resistance of the second source follower circuit is connected to the cathode of the parasitic diode, and a common bias voltage is applied to the gate of the FET of the first source follower circuit and the gate of the FET of the second source follower circuit. The voltage between the anode and the cathode of the parasitic diode is converted into a difference between a current flowing through the first source follower circuit and a current flowing through the second source follower circuit. Power supply.
8. The power supply device according to claim 6,
   A voltage detection circuit for detecting a voltage between the anode and cathode of the parasitic diode;
   The voltage detection circuit includes a first circuit and a second circuit,
   The first circuit includes a diode connected in series to the cathode of the parasitic diode, a first resistor, a first PMOSFET, a second resistor, and a first NMOSFET, and the gate of the first PMOSFET. And the drain are short-circuited, the gate and drain of the first NMOSFET are short-circuited, and the source is grounded,
   The second circuit includes a diode connected in series to the anode of the parasitic diode, a third resistor, a second PMOSFET, a fourth resistor, and a second NMOSFET,
   The gate of the second PMOSFET is connected to the gate of the first PMOSFET, the source of the second NMOSFET is grounded, and is connected to the gate of the first NMOSFET;
   The second PMOSFET has the same characteristics as the first PMOSFET, the second NMOSFET has the same characteristics as the first NMOSFET,
   The resistance value of the fourth resistor is set to be smaller than the resistance value of the second resistor, and the resistance value of the third resistor is equal to or greater than the resistance value of the first resistor. Is set to
   The power supply apparatus, wherein the drain voltage of the second NMOFET is configured to change according to a voltage between an anode and a cathode of the parasitic diode.

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