WO1999034648A1

WO1999034648A1 - Electronic lamp ballast

Info

Publication number: WO1999034648A1
Application number: PCT/EP1998/007426
Authority: WO
Inventors: Norbert Primisser; Reinhard BÖCKLE; Stefan Koch; Stefan Rhyner
Original assignee: Tridonic Bauelemente Gmbh
Priority date: 1997-12-23
Filing date: 1998-11-19
Publication date: 1999-07-08
Also published as: ES2195438T3; EP1048190A1; AU1962499A; DE59808310D1; EP1048190B1; ZA9811774B; ATE240030T1

Abstract

The invention relates to an electronic lamp ballast for operating at least one gas-discharge lamp (G1, G2). The electronic lamp ballast comprises a control circuit (IC2), which monitors a voltage potential applied to the input connection (NP) and, depending on the value of the voltage potential, operates the gas-discharge lamp (G1, G2) with or without preheating. The voltage potential monitored by the control circuit (IC2) can in particular depend on the charge status of an energy storage circuit (R22, C17) which is charged while the gas-discharge lamp (G1, G2) is operating and is discharged at a certain time constant when the electronic lamp ballast is switched off. This allows for dynamic switching between cold-start and warm-start operations.

Description

Electronic ballast

The present invention relates to an electronic ballast for operating at least one fluorescent or gas discharge lamp according to the preamble of claim 1.

Such an electronic ballast is known for example from EP-Bl-0338 109. Fig. 10 shows the basic structure of this electronic ballast.

The electronic ballast shown in FIG. 10 first comprises a circuit A which is connected to the AC network. This circuit A serves as an RF harmonic filter for reducing the harmonic harmonics of the mains frequency and for radio interference suppression.

The circuit A is followed by a rectifier circuit B, which converts the mains voltage into a rectified intermediate voltage and supplies it to an inverter circuit D via a harmonic filter C, which serves to smooth the intermediate voltage. This inverter D serves as a controllable AC voltage source and converts the DC voltage of the rectifier B into a variable AC voltage. The inverter D generally comprises two controllable switches (not shown), for example MOS field-effect transistors. The two switches are connected in the form of a half-bridge circuit and are alternately controlled with the aid of a corresponding bridge driver such that one of the switches is switched on and the other is switched off. The two inverter switches are connected in series between a supply voltage and ground, a load circuit or output circuit E in which a gas discharge lamp or fluorescent lamp G is arranged being connected to the common node between the two inverter switches. This output circuit E comprises a series resonance circuit, via which the “chopped” high-frequency AC voltage of the inverter D is fed to the fluorescent lamp G.

Before the ignition voltage is applied to the fluorescent lamp G, the lamp electrodes of the fluorescent lamp G are preheated in order to extend the life of the lamp. The preheating can be carried out, for example, with the aid of a heating transformer, the primary winding of which is connected to the series resonant circuit, while the secondary windings of the heating transformer are coupled to the individual lamp filaments. In this way it is possible to supply the lamp filaments with energy even in the ignited mode. In preheating mode, the frequency is that of the inverter D changed AC voltage with respect to the resonance frequency of the series resonance circuit of the output circuit E changed such that the voltage applied to the gas discharge lamp G does not cause the lamp to ignite. In this case, an essentially constant current flows through the lamp electrodes of the lamp, which are designed as filaments, whereby the lamp filaments are preheated. After the preheating phase, the frequency of the alternating voltage supplied by the inverter D is shifted close to the resonant frequency of the series resonant circuit, as a result of which the voltage applied to the gas discharge lamp G increases, so that the gas discharge lamp G is ignited.

During the preheating, ignition and operation of the gas discharge lamp G, certain faults can occur which have to be recognized in order to be able to react accordingly. For this purpose, the electronic ballast has a control circuit F which monitors various circuit sizes of the electronic ballast and generates a corresponding control signal for the inverter D when a limit value is exceeded in order to change the frequency of the alternating voltage generated by the inverter D depending on the detected fault . For example, the control circuit F can monitor the lamp voltage, the preheating voltage, the lamp operating current, the impedance phase angle of the output circuit E or the DC voltage generated by the rectifier B and set the inverter frequency such that the lamp voltage, the preheating voltage or the lamp current have a predetermined limit value do not exceed, the direct current power taken from the rectifier B is as constant as possible or a capacitive operation of the series resonance output circuit E is avoided.

As has been described above, it is known to preheat the lamp filaments of the gas discharge lamp with the aid of a preheating operation in order to extend their life. However, this leads to the fact that a longer period of time elapses as a result of the preheating operation until the gas discharge lamp can be ignited and put into operation. However, cases are conceivable in which it is possible to dispense with the preheating of the lamp electrodes, for example if the corresponding gas discharge lamp has already been operated shortly beforehand.

The present invention is therefore based on the object of providing an electronic ballast which enables the simplest possible switchover between a cold start operation without preheating the lamp filaments and a warm start operation with preheating the lamp filaments. This object is achieved according to the present invention by an electronic ballast with the features of claim 1. The subclaims describe advantageous and preferred exemplary embodiments of the present invention.

According to the present invention, the electronic ballast has a control circuit which monitors a circuit or operating variable of the electronic ballast and, depending on the value of this circuit variable, operates the electronic ballast either with a preheating mode or without a preheating mode. The circuit size monitored by the control circuit is advantageously a voltage potential present at a corresponding connection, the connection being able to be connected optionally to a high or a low supply voltage potential (e.g. via a resistor). Depending on which supply voltage potential the resistor is connected to, the control circuit thus determines whether a warm start or cold start operation is desired. The application of the high or low supply voltage potential to the connection mentioned above can be carried out, for example, simply by changing the resistance described above by a user.

According to a preferred embodiment of the invention, the electronic ballast is designed such that it automatically detects when the electronic ballast is restarted or restarted whether the correspondingly controlled gas discharge lamp has already been in operation, so that the lamp electrodes of the gas discharge lamp may not be preheated can be. This takes place with the aid of an energy storage circuit which is charged during the operation of the gas discharge lamp and is discharged with a specific time constant after the electronic ballast has been switched off. The control circuit provided according to the invention automatically detects the charge still remaining in the energy storage circuit when the electronic ballast is restarted, the charge recorded being a measure of the switch-off time of the electronic ballast and thus also of the temperature of the lamp electrodes still present. If the charge remaining in the energy storage circuit when the electronic ballast is switched on is still sufficiently high, the control circuit controls the gas discharge lamp without preheating operation. Otherwise a preheating operation is carried out. The energy storage circuit can in particular be formed by an RC element connected to the control circuit. With the help of the solution according to the invention it can be ensured that the gas discharge lamp can be put into operation as quickly as possible by dispensing with an unnecessary preheating operation. This measure also protects the lamp electrodes of the gas discharge lamp, since excessive heating of the lamp electrodes is also prevented. This ensures that even when the gas discharge lamp is started up without preheating operation, the lamp electrodes have a sufficient temperature to ignite the gas discharge lamp.

The invention is explained in more detail below on the basis of preferred exemplary embodiments with reference to the attached drawing.

1 shows a circuit diagram of a preferred exemplary embodiment of an electronic ballast according to the invention,

FIG. 2 shows an enlarged illustration of a control circuit shown in FIG. 1 with a corresponding external sound system of this control circuit,

3 shows a block diagram of the control circuit shown in FIG. 2,

4 shows a circuit diagram of a current detection block shown in FIG. 3,

5a and 5b show illustrations for explaining the capacitance current detection with the aid of the current detection block shown in FIGS. 3 and 4,

6 shows a circuit diagram of a voltage detection block shown in FIG. 3,

7a and 7b show illustrations for explaining the lamp change detection with the aid of the voltage detection block shown in FIGS. 3 and 6,

8a shows a circuit diagram to explain the warm / cold start mode switchover according to a first exemplary embodiment of the present invention,

8b shows a circuit diagram to explain a dynamic one

Warm / cold start mode switching according to a second exemplary embodiment of the present invention,

9 shows an example of various operating states controlled by the electronic ballast according to the invention, Fig. 10 shows a block diagram of a known electronic ballast, and

11a to 11d show illustrations for explaining a further function of the voltage detection block shown in FIGS. 3 and 6.

The electronic ballast shown in FIG. 1 first comprises a circuit A, which is connected on the input side to a supply voltage, for example a mains voltage, and is used for radio interference suppression. The circuit A is constructed in the usual way and includes, for example, capacitive input filters and, if necessary, harmonic chokes. In Fig. 1, a capacitor C2 and a symmetry transformer L1 are shown only by way of example, it being possible for a surge arrester or a VDR with the designation F1 to be connected in parallel.

The circuit B following the circuit A comprises a full-wave rectifier bridge with diodes VI-V4. The rectifier circuit B converts the supply AC voltage present on the input side into a rectified intermediate voltage. The rectifier circuit B can therefore be omitted if the electronic ballast is operated with direct voltage.

The following circuit part C serves for harmonic filtering and smoothing the intermediate voltage supplied by the rectifier B. The circuit C shown in FIG. 1 comprises, for example, capacitors C3, C1, a diode V5, a coil L2, a MOS field-effect transistor T1 and a control circuit IC1 designed as an integrated circuit. The control circuit IC1 is connected to a supply voltage potential VCC and can be connected to the other circuit elements in such a way that it receives different voltage potentials U or currents I. The structure of the circuit C shown in FIG. 1 is of course to be understood purely by way of example.

An inverter circuit D is driven by the harmonic filter C shown in FIG. 1, the essential elements of which are two controllable switches connected in series between a supply voltage line and ground, in the present example in the form of MOS field-effect transistors T2 and T3. The two inverter switches T2, T3 are connected to form a half bridge and are each controlled, ie opened and closed, with the aid of a control circuit IC2 designed as an integrated circuit. The control circuit IC2 thus also takes on the function of a bridge driver and is connected to or coupled to a corresponding supply voltage line VCC. The Inverter circuit D generates an alternating voltage with variable frequency and / or duty cycle depending on the rectified intermediate voltage generated by rectifier circuit B. In general, the inverter D is constructed in the usual way and its function is sufficiently known, so that a further explanation can be dispensed with here. It is only important at this point that the control circuit IC2 controls the two inverter switches T2 and T3 alternately, depending on the control signals supplied, so that a "chopped", high-frequency AC voltage occurs at the connection point between the two inverter switches T2 and T3.

A series resonance output circuit or load circuit E is connected to the inverter D. In the present case, the load circuit E is designed for the connection of two gas discharge lamps Gl, G2 in a tandem configuration. Of course, the load circuit E can also be modified such that only one gas discharge lamp or more than two gas discharge lamps can be operated. From Fig. 1 it can be seen that the load circuit E has a series resonance circuit consisting of a resonance circuit coil L3 and a resonance circuit capacitor C14. This series resonant circuit or the resonant circuit coil L3 is connected to the connection point between the two inverter switches T2 and T3 and the resonant circuit capacitor C14 is arranged such that it is connected in parallel with the gas discharge lamp or gas discharge lamps Gl, G2 to be operated. The high-frequency AC voltage generated by the inverter D is supplied to the gas discharge lamps Gl and G2 via the series resonance circuit.

As has already been explained, according to the example shown in FIG. 1, the two gas discharge lamps Gl and G2 are connected in a tandem configuration to the load circuit E or the electronic ballast. This means that - as can be seen from Fig. 1 - the upper filament of the upper gas discharge lamp Gl and the lower filament of the lower gas discharge lamp G2 are connected directly to the load circuit E, while the lower filament of the upper gas discharge lamp Gl and the upper filament of the lower Gas discharge lamp G2 connected to each other and connected to the load circuit E. Before the ignition voltage is applied to the gas discharge lamps G1, G2, these are preheated in order to extend the life of the gas discharge lamps. For this purpose, a heat exchanger L4 is provided according to FIG. 1, the primary winding of which is connected in series with the resonant circuit capacitor C14, while the secondary winding connects the lower filament of the upper gas discharge lamp Gl and the upper filament of the lower gas discharge lamp G2 to one another. In the preheating mode, the frequency of the AC voltage supplied by the inverter E is related the resonance frequency of the series resonance circuit is set such that the voltage across the resonance circuit capacitor C14 and thus across the gas discharge lamps Gl and G2 does not cause the gas discharge lamps to ignite. In this case, an essentially constant preheating current flows through the filaments of the gas discharge lamps Gl, G2. The capacitor C15 shown in FIG. 1 adapts the preheating voltage in the tandem configuration of the gas discharge lamps Gl and G2 shown in FIG. 1. The previously explained principle of preheating can of course also be transferred in a simple manner to the operation of a gas discharge lamp or more than two gas discharge lamps. In addition to the tandem configuration, a parallel configuration or parallel connection of a plurality of gas discharge lamps Gl, G2 is also conceivable. In Fig. 1, however, the tandem configuration of the gas discharge lamps Gl, G2 is shown, since in such a lamp configuration with the help of the electronic ballast shown in Fig. 1, a lamp change of both the upper and the lower gas discharge lamp can advantageously be determined in a simple manner. The lamp change detection is explained in more detail below. Resistor R12 shown in FIG. 1 is also used for the purpose of detecting lamp changes.

After the preheating phase has elapsed, the frequency of the alternating voltage supplied by the inverter D is shifted into the vicinity of the resonant frequency of the series resonant circuit via the control circuit IC2, as a result of which the voltage across the resonant circuit capacitor C14 and the gas discharge lamps Gl, G2 is increased, as a result of which these gas discharge lamps are ignited.

After the gas discharge lamps have been ignited, the electronic ballast shown in FIG. 1 goes into the actual operating phase, in which the frequency of the alternating voltage supplied by the inverter D is continuously set, for example, in such a way that the most constant lamp current flows or starts through the gas discharge lamps Gl, G2 the gas discharge lamps have as constant a lamp voltage as possible. As will be explained in more detail below, the electronic ballast shown in FIG. 1 has a number of fault detectors which monitor certain circuit sizes of the electronic ballast, in particular the load circuit E, and trigger a corresponding control of the inverter D when a specific fault is detected For example, to avoid the occurrence of an overvoltage on the gas discharge lamps G2 and G2, a rectification effect in the gas discharge lamps G1, G2 or a capacitive operation of the load circuit E. To control the inverter D, a circuit module is used which, at the heart, comprises the control circuit IC2 already mentioned and several external components as external circuitry for the control circuit IC2. The main external components are six resistors RIO, R13 - R16 and R21, R22 and two capacitors C7 and C17. As shown in Fig. 1, the individual external components are connected to respective input terminals of the control circuit IC2. The external components connected to the control circuit IC2 serve primarily to detect certain circuit sizes of the electronic ballast, so that these can be evaluated in the control circuit IC2.

FIG. 2 shows an enlarged representation of the control circuit IC2 shown in FIG. 1 and the external wiring of the individual input connections of the control circuit IC2. Only the essential connections and external components are shown in FIG. 2. In the present example, the control circuit IC2 is advantageously designed as an application-specific integrated circuit (Application Specific Integrated Circuit, ASIC) and accommodated in a multi-pole SMD housing (Surface Mounted Device). In the present case, the control circuit IC2 is suitable both for the operation of a single lamp output circuit E and for the operation of a load circuit E designed for a tandem configuration shown in FIG. 1 with a plurality of gas discharge lamps.

As has already been mentioned and can be seen in particular from FIG. 2, the control circuit IC2 has a plurality of connections which have the following functions. The reference potential, ie the ground potential, for the individual analog and digital function blocks of the control circuit IC2 is applied to the connection GND. From Fig. 1 it can be seen that the ground potential of the entire electronic ballast is grounded via a coupling capacitor C1. The internally generated supply voltage for the individual analog and digital function blocks of the control circuit IC2 is provided at the connection VDD, which is connected to the ground potential via the coupling capacitor C7 (cf. FIG. 1). The connection NP serves, as will be explained in more detail below, for the external setting and detection of the preheating method, ie for the selection between a cold start and warm start operation. In particular, the NP connection is connected externally in such a way that dynamic selection of the preheating method is possible. The terminal VL1 detects the divided lamp voltage of the gas discharge lamps Gl, G2 via the resistors RIO and R14, R15 shown in FIG. 1 and partly in FIG. 2 and thus serves primarily for monitoring the lamp voltage. Analogously, the connection ILC is used to monitor the resistors R13 and R16 shown in FIG. 1 and partly in FIG. 2 Output circuit or load circuit current (choke current) or for monitoring the lamp current flowing through the gas discharge lamps Gl, G2 after their ignition, by using the shunt resistor R16 to detect a voltage proportional thereto and to supply the control circuit via the connection ILC. The VLI connection is thus used for voltage monitoring, while the ILC connection is used for current monitoring. The two output connections OUTL and OUTH serve to control the low-lying or high-lying half-bridge switch T3 and T2 shown in FIG. 1. For this purpose, control signals (TTL level) for switching the two inverter switches T2 and T3 on and off are provided at the output connections OUTL and OUTH. The connection VCC of the control circuit IC2 is finally the central supply voltage connection of the control circuit IC2.

The supply voltage range can include 10-18V, for example. Furthermore, the control circuit IC2 controls the inverter switches T2 and T3 in such a way that an alternating voltage of variable frequency with an operating frequency range of, for example, 40-80 kHz is generated on the output side by the inverter circuit D.

The control circuit IC2 forms the heart of the entire electronic ballast shown in FIG. 1 and accordingly comprises a large number of different functions. For example, the control circuit IC2 can be used to dynamically determine the preheating method for the connected gas discharge lamp (s) and to switch between a cold start and a warm start operation. For this purpose, the control circuit IC2 ensures a defined preheating operation with a defined preheating time and a defined preheating current. The control circuit IC2 likewise ensures a predefined ignition operation with a defined ignition time and a defined ignition voltage. For example, the preheating current and the lamp operating current or the lamp voltage can be detected via the connections ILC or VLI of the control circuit IC2 and regulated to a value that is as constant as possible. Furthermore, the control circuit IC2 monitors the capacitive operation of the load circuit E via the current connection ILC. The occurrence of a constant light effect in a connected gas discharge lamp Gl, G2 can also be detected via the voltage connection VLI. Likewise, the occurrence of a gas defect, which leads to an overvoltage in the corresponding gas discharge lamp, can be detected with the aid of the voltage connection VLI and the electronic ballast can accordingly be switched off in this case. A special function of the control circuit IC2 is the detection of a lamp change, the lamp change detection in the tandem configuration shown in FIG. 1 being in particular independent of the changed lamp, ie both the upper gas discharge lamp Gl and the lower one can be changed Gas discharge lamp G2 can be detected. Finally, a (preferably digitally implemented) sequence control is implemented in the control circuit IC2, which ensures that the gas discharge lamp (s) connected to the electronic ballast are controlled in accordance with predetermined operating states, whereby from one operating state to a new operating state only when fulfilled at least one specific condition can be changed. Within each operating state, operating state-dependent monitoring of certain variables of the electronic ballast is possible, so that different error variables can be monitored and evaluated differently depending on the respective operating state. With regard to the errors, an event-filtered error evaluation takes place in particular, ie with the help of digital event filters, for example, it is ensured that the existence of an error is only concluded if the corresponding error actually occurs several times in succession.

In addition to the functions briefly summarized above, the control circuit IC2 has further functions, all of which will be explained in more detail below with reference to the accompanying drawing.

Fig. 3 shows a block diagram of the internal structure of the control circuit IC2 described above. First, a module 100 is coupled to the current connection ILC, which is used, among other things, for the previously described current detection and capacitive current detection of the load circuit. The evaluation of the current sensed via the connection ILC is carried out in particular with the aid of a regulator formed by a comparator circuit. In order to keep the circuit complexity really low, this comparator circuit is also supplied and evaluated with the voltage signal received by the voltage connection VLI of the control circuit IC2 and processed by a module 200. The module 200 is used in particular for detecting the lamp voltage, for rectifying effect detection and for lamp change detection. Furthermore, a further module 300 is coupled to the connection NP, which module is used to detect the warm or cold start operation when preheating the gas discharge lamp (s) to be controlled and to implement a dynamic preheating operation. A voltage regulator module 400 is connected to the supply voltage connections VCC and VDD, which has an internal voltage regulator which provides a regulated, very precise voltage for the voltage supply of all internal function blocks. Another module 500 serves as a source for all required reference quantities, ie reference voltages and reference currents, in the control circuit IC2. An oscillator 600 serves as an internal clock generator of the control circuit IC2, a time base generator 700 coupled to it depending on the predetermined clock of the oscillator 600 internal time variables for the sequence control of the Control circuit IC2, such as the preheating or ignition time, derived. Another module 800 is used to implement the sequence control of the individual operating states of the entire electronic ballast and interacts closely with another module 900, which is used for measuring phase control. The module 900 is used in particular for event-filtered evaluation of certain error quantities of the electronic ballast and for the measurement phase-dependent control of all switches of the individual function blocks of the control circuit IC2. The sequence controller 800 evaluates the event-filtered status messages of the measurement phase controller 900 and controls the individual operating states of the electronic ballast or the control circuit IC2 depending on the time variables specified by the time base generator 700. Finally, the control circuit IC2 has a further module 1000 for controlling the inverter. With the aid of this module 1000, frequency setting signals supplied by the measuring phase control 900 are converted into corresponding control signals for the upper inverter switch (via the output connection OUTH) or the lower inverter switch (via the output connection OUTL).

The control circuit IC2 can comprise both analog and digitally implemented function blocks. In the present case, the digital part of the control circuit IC2 designed as ASIC comprises the time base generator 700, the sequence control 800, the measurement phase control 900 and the inverter control 1000. In particular, the control circuit IC2 can be equipped in such a way that the digital part corresponds to the analog part in terms of the area required by the control circuit IC2.

FIG. 4 shows a detailed circuit diagram of the current detection module 100 shown in FIG. 3. FIG. 4 also shows the resistors R13 and R16 which are connected externally to the current connection ILC of the control circuit and which are also shown in FIG. 1.

Internally, a reference current Irefl is added to the signal detected at the current connection ILC in order to ensure that the signal to be processed by the current detection module 100 is always in the operating voltage range of the control circuit.

As shown in FIG. 4, an integrator circuit 105 is provided, which is used to integrate the input signal supplied to it. The entire function block 105 is implemented in such a way that the integrator function can be used both for measuring the lamp current (via the ILC connection) in normal operation and for identifying rectification effects (via the VLI connection). The integrator circuit 105 can have sample and hold elements, which sample the input signal of the integrator alternately every period of the internal clock generator (cf. module 600 in FIG. 3). The charge thereby stored in the sample and hold elements is passed on to an integration amplifier of the integrator circuit 105. This process is repeated cyclically.

The integrator 105 can have an internal controllable switch which bridges the aforementioned sample and hold elements and is closed during the duration of the offset adjustment of the integrator 105. In this way, any signal, in particular the signal present at the input connection ILC, can be applied to the actual integration amplifier via the switch S105 or a reference voltage potential for rectifying effect detection from the voltage block 200 via the switch S107 during the initialization phase.

The actual integration amplifier of integrator 105 has the task of integrating the current measurement signal at the ILC connection in a time-controlled manner. In the event that the current measurement signal present at the ILC connection is integrated by the integration amplifier of the integrator circuit 105, the switch S105 is closed, while in the case of the rectification effect evaluation the reference potential for the rectification effect evaluation supplied via the switch S107 is applied to the integrator circuit 105.

Finally, a comparator 103 serves as the actual controller, which carries out the required setpoint / actual value comparison and is connected to the output of integrators 105. The arrangement of this comparator 103 shown in FIG. 4 makes it possible to use the comparator 103 very flexibly. By actuating a switch S124 accordingly, different reference voltages or reference values can be connected or applied to the comparator 103, reference voltages Vrefl-Vrefό being shown by way of example in FIG. The reference potential Vrefl and Vref2 corresponds, for example, to a desired preheating voltage during a preheating operating state. During the preheating operation, the reference voltage Vrefl or Vref2 is thus applied to the comparator 103 with the aid of the controllable switch S124, so that the measurement signal which is present at the ILC connection and is not integrated is compared with the reference value Vrefl or Vref2 respectively applied. In normal operation, for example, the reference potential Vref3 corresponds to the integration start value of the integration amplifier of the integrator 105, so that when this reference potential Vref3 is applied, the comparator 103 can detect the actual change in the integration result. The reference potentials Vref4 or Vrefό can correspond to a positive or negative limit value for the lamp voltage of the connection VLI supplied and integrated via the switch S107, in order to be able to reliably detect the occurrence of a rectification effect by comparison with these two limit values if the integration result is exceeded in a positive or negative direction. For this purpose, the further reference potential Vref5 is also used, which is added during the rectification effect detection and corresponds to the output or start value for the integration of the lamp voltage supplied via the switch S107. By taking into account the start values of the integration amplifier of the integrator 105 given by the reference potentials Vref3 or Vref5, the change in the corresponding integration variable that is actually present relative to the corresponding start value can thus be determined with the aid of the comparator 103.

The output signal of the comparator 103 is fed to the measuring phase controller 900 shown in FIG. 3, which evaluates it and evaluates it differently depending on the current measuring phase. This is how the measuring phase control ensures

900 for example for a corresponding adjustment of the output frequency of the

Inverter of the electronic ballast, if the current measurement signal monitored by the comparator 103 of the connection ILC deviates from the predetermined target value Vref3. In the case of the rectification effect evaluation, however, as will be explained in more detail below, the measurement phase control generates an event-filtered signal which indicates whether there is a rectification effect in a connected gas discharge lamp or not. This signal is evaluated by the sequence control block 800 shown in FIG. 3 and used to control the operating state of the entire electronic ballast.

In the present case, it is proposed in particular to regulate the peak value of the preheating current I _HZ during the preheating operation. For normal operation, however, it is proposed to regulate the mean value or the effective value of the lamp operating current I _L.

As shown in FIG. 4, the measurement signal present at the connection ILC can also be monitored and evaluated bypassing the integrator circuit 105 in order, for example, to detect a capacitive operation of the load circuit of the electronic ballast. For this purpose, a detector can be provided for detecting a capacitive current flowing in the load circuit, which, for example, determines the phase angle of the load circuit, ie the phase difference between the load circuit voltage and the load circuit current (Capacitive current detection). The result of this monitoring or evaluation can also be fed to the measuring phase controller 900.

The occurrence of a capacitive current in the load circuit will be explained in more detail below with reference to FIGS. 5a and 5b.

FIG. 5a shows an enlarged illustration of the essential elements of the inverter D already shown in FIG. 1 and of the load circuit E. For the sake of simplicity, FIG. 5a assumes that only one gas discharge lamp Gl is connected to the load circuit. 5a shows the two inverter switches T2 and T3 connected in series. As already shown in Fig. 1, the load circuit with its series resonance circuit is connected to the connection point between the two inverter switches T2 and T3, i.e. the resonance circuit coil L3 is connected in parallel with the resonance circuit capacitor C14 to the lower inverter switch T3. The resonant circuit capacitor C14 is also connected in parallel to the gas discharge lamp Gl. Free-wheeling diodes VI 1 and V12 are connected in parallel to the individual inverter switches T2 and T3 and serve to protect the respective inverter switch.

5b shows on the one hand the switch-on states of the two inverter switches T2 and T3 and the current profile of the current I _u flowing through the choke L3 and the time profile of the voltage potential V _L occurring at the connection point between the two inverter switches T2 and T3. At the time the upper or lower inverter switch T2 or T3 is switched on, a current flows in the freewheeling diode of the inverter switch to be switched on and the inverter half bridge switches the resonant load circuit inductively, ie the voltage or potential V _{L leads} the choke current I _u . In contrast to this is the capacitive switching of the resonance load of the resonance load circuit. In this working area, a current flows in the freewheeling diode opposite the inverter switch to be switched on, as a result of which a high reverse voltage occurs at the forward-current-carrying freewheeling diode of one of the two inverter switches. This in turn follows to very high recovery currents, as a result of which high switching losses occur in both inverter switches T2 and T3.

FIG. 5a shows the course of the individual currents Ij-I ₄ , which occur during the time intervals t ,, - t ₄ shown in FIG. 5b in the case of an inductive or capacitive inductor current 1 ^. The above-mentioned phenomenon occurs in particular in the case of load circuit voltages V _L with an output frequency close to the resonance frequency of the series resonance circuit, which is the case in particular when the gas discharge lamp Gl is ignited, an inductive current initially flowing in the load circuit, which leads to heating of the coil L3 leads. Due to the heating of the coil L3, its inductance drops, so that a transition suddenly occurs from the inductive area to the faulty capacitive area.

To detect the undesired capacitive operation of the load circuit, the level of the current amplitude of the load circuit detected via the input ILC can now be monitored and compared with a fixed, predetermined reference value. The magnitude of the current amplitude is advantageously detected at the time when the lower inverter switch T3 is switched on, since in this case the polarities of the measured values to be recorded are favorable for processing within the control circuit IC2 designed as an ASIC. If the detected current value is below the limit value specified by the corresponding reference potential, the presence of capacitive operation is inferred from the load circuit, and an output signal with a high level can be generated which is evaluated by the measurement phase control block 900 shown in FIG. 3 and finally by the inverter control block 1000, also shown in FIG. 3, is converted into control signals for the two inverter switches T2 and T3 in such a way that they are alternately switched on and off at an increased frequency in order to increase the working frequency and thus counteract the capacitive operation.

The structure and the function of the voltage detection block 200 already indicated in FIG. 3, which evaluates or processes the measurement signals present at the voltage connection VLI, will be explained in more detail below with reference to FIG. 6.

6 shows on the one hand the internal structure of the voltage detection block 200 and the external circuitry of the control circuit coupled to the connection VLI of the voltage detection block 200. In particular, in accordance with FIG. 1, it can be seen from FIG. 6 that a series resistor RIO is coupled on the one hand to the connection VLI and on the other hand to a voltage divider consisting of resistors R14 and R15, the two voltage divider resistors R14 and R15 being connected in parallel to the gas discharge lamp Gl and are connected to the gas discharge lamps Gl and G2 connected in tandem in FIG. 1. For the sake of simplicity, it is assumed in FIG. 6 that, in contrast to FIG. 1, only one gas discharge lamp G1 is driven, to which the resonant circuit capacitor C14 is also connected in parallel. The two resistors R14 and R15 have the task of dividing down the voltage applied to the gas discharge lamp Gl so that a measuring signal representative of the lamp voltage can be fed to the voltage terminal VLI of the voltage detection block 200 with the aid of the resistor RIO acting at the connection point between the resistors R14 and R15 .

Advantageously, the three external resistors RIO, R14 and R15 are variable, so that - analogously to the power connection ILC (cf. resistors R13, R16) - a connection of the control circuit completely independent of one another at different times includes a total of three different control variables of the electronic ballast Can be set or controlled using one and the same controller. By setting the resistance values of the resistors RIO, R14 and R15, the setpoints for the control of the three different control variables can be set or predefined depending on the lamp type or the electronic ballast type currently being used. In the present case, the following variables of the electronic ballast can be set using the three external variable resistors RIO, R14 and R15: the maximum lamp voltage positive / negative, the amplitude of the AC voltage component of the lamp voltage signal and the signal increase of the lamp voltage signal for rectification effect evaluation.

As can be seen in FIG. 6, an internal reference current source is again provided, which applies an additional internal current Iref2 to the measurement signal present at the voltage connection VLI. In contrast to the ILC connection shown in Fig. 4, however, in this case the reference current Iref2 is activated with the help of the controllable switch S207 only during the evaluation of the rectification effect, i.e. closed. All other evaluations connected to the VLI connection relate to the signal present at the VLI connection without an additional reference current Iref2, i.e. without DC offset. Accordingly, all other detectors on the VL1 connection are deactivated during the rectification effect evaluation, since they would otherwise give incorrect results.

By connecting the reference current Iref2, the signal present at the connection VLI is raised again. Analogously to feeding in the reference current Irefl shown in FIG. 4 at the current connection ILC, this is harmless in the rectification effect evaluation, since - as will be explained in more detail below - the rectification effect detection by evaluating the output signal of the in FIG. 4 Integrator circuit shown is carried out, the DC component Irefl or Iref2 is eliminated due to the averaging by the integrator circuit.

First, the rectification effect detection is to be explained in more detail with the help of the present control circuit. As with other lamps, gas discharge lamps, due to the wear of the heating filament, have the effect at the end of the life of the gas discharge lamps that the lamp electrodes wear out unevenly over time, i.e. the removal of the emission layers on the lamp electrodes is different. Due to the different wear of the lamp electrodes, there are differences in the emissivity of the two lamp electrodes. The result of this is that a higher current flows from one lamp electrode to the other when the corresponding gas discharge lamp is operated than vice versa. The time course of the lamp current thus shows an increase in a half-wave. The different removal of the two lamp electrodes thus creates asymmetries which not only lead to a stronger flicker of light at the end of the life of the gas discharge lamp, but even in extreme cases only permit the gas discharge lamp to be operated during one half-wave. In this case, the gas discharge lamp acts like a rectifier, so that the effect described above is referred to as the "rectifying effect".

The rectification effect explained above also has the consequence that the more worn electrode, which has a higher work function than the other electrode, heats up more than the other electrode when the gas discharge lamp is started up. The work function is generally the minimum energy required to detach an electron from a metal, in the present case from a lamp electrode. The heating of the lamp electrode described above can become so strong, in particular in the case of lamps with a small diameter, that parts of the lamp glass bulb can melt.

Therefore, with the help of the present control circuit, each controlled lamp is monitored for the occurrence of a rectification effect, so that a reaction can be made accordingly when a rectification effect is detected.

As has already been indicated above, the actual rectification effect detection does not take place in the voltage detection block 200 shown in FIG. 6, but in the current detection block 100, since the integrator circuit of the current detection block 100 and the downstream comparator 103 (cf. FIG. 4) are also used for the rectification effect detection . In this way, the number of monitoring of the electronic ballast or the gas discharge lamp (s) required components can be reduced.

In the voltage detection block 200 shown in FIG. 6, only the signal conditioning of the measurement signal present at the connection VLI takes place. For this purpose, switch S207 is first closed in order to raise the AC voltage signal present at connection VLI positively. At the downstream coupling capacitor C201, however, only the AC voltage component of the measurement signal thus processed can pass. Therefore, the signal must be raised again after the coupling capacitor C201, which takes place after a certain settling time with respect to the current source Iref2 by closing a switch S201. To raise the measurement signal again, an internally defined reference voltage VrefS, which is not influenced from outside, is used, which is therefore known to the entire control circuit. In the case of the rectification effect evaluation, this internal reference voltage Vrefδ - as has already been explained with reference to FIG. 4 - is also present at the integration amplifier of the integrator circuit 105.

The switch S207 shown in FIG. 6 is advantageously closed some time before the expected zero crossing of the lamp voltage signal present at the connection VLI, so that transient processes caused by the capacitor C201 cannot additionally falsify the measurement signal. Switch S201 is opened again exactly at the calculated zero crossing of the lamp voltage. The signal present at the switch S107 shown in FIGS. 4 and 6 corresponds at this point in time to the AC voltage amplitude at the connection VLI, while the DC component of the signal present at the switch S107 corresponds to the reference voltage VrefS that is switched on. By closing the switch S107, the measurement signal of the connection VLI prepared in this way is finally fed to the integrator circuit 105 shown in FIG. 4, as has already been explained above. The switching state of the switch S107, like all other controllable switches of the entire control circuit IC2, is controlled by the measuring phase controller 900 shown in FIG. 3. The individual switches shown in FIG. 4 are closed or opened by the measuring phase controller 900 in such a way that, with the aid of the comparator 103, an averaged evaluation of the current measuring signal present at the terminal ILC or of the voltage measuring signal present at the terminal VLI is possible via the upstream integrator circuit. Furthermore, by corresponding actuation of the controllable switches of the current detection block 100 shown in FIG. 4, the comparator 103 can also be connected directly to the current measurement connection ILC, bypassing the integrator circuit, in order thus to evaluate or regulate the peak value of the current measurement signal at the connection ILC. As has already been explained predetermined by the measuring phase control 900, which of the measuring or. Control states are assumed.

The rectification effect detection principle implemented in the present control circuit IC2 provides that the lamp voltage detected via the voltage connection VLI is integrated with the aid of the integrator circuit of the current detection block 100 shown in FIG. 4, and then the deviation from a predetermined desired value is evaluated. In particular, the measurement signal corresponding to the lamp voltage is integrated over a full period or a multiple of a full period of the lamp voltage, and then the deviation of the integration result from the original integration start value is evaluated. For this purpose, the comparator 103 is supplied with the integration start value by applying the corresponding reference potential Vref5. With the help of the switch S124, the comparator 103 can also be given a positive limit value or a negative limit value for the rectification effect detection in the form of the further reference potentials Vref4 or Vrefό. The potential Vref5 can be, for example, 3.0V, while a value of 4.0V can be used as the positive reference potential Vref4 and a value of 2.0V can be used as the negative reference potential Vrefό.

With the aid of the evaluation circuit shown in FIG. 4, it is possible to recognize the rectifying effect both in the positive and in the negative direction. The output signal of the comparator shown in FIG. 4 is in turn fed to the measurement phase controller 900 which, after detection of a rectification effect, outputs a corresponding status message or error message to the sequence controller 800 shown in FIG. 3. However, since a rectification effect should not be prematurely concluded if, for example, it only occurs for a short time, the measurement phase controller 900 carries out an event-filtered revision of this error message and ensures that an error message indicating the rectification effect is only output to the sequence controller 800 if the rectification effect occurs continuously over a long period of time. In principle, this applies not only to the error message indicating a rectification effect, but also to all error or status messages output by the measuring phase controller 900 to the sequence controller 800. However, the aforementioned revision of the rectification effect error message is particularly useful because the rectification effect is a creeping effect that occurs with a time delay. Therefore, the measuring phase controller 900 only outputs a rectification effect error message to the sequence controller 800 if a rectification effect is detected 32 times in succession every 255th period of the lamp voltage by the comparator 103 shown in FIG. 4. As soon as no rectification effect has been detected during a period of the lamp voltage, the counter of the measuring phase control 900 assigned to the rectification effect is reset to zero and the evaluation of the rectification effect error signal of the comparator 103 is started again.

As will be explained in more detail below, the occurrence of a rectification effect is only taken into account in the operating state of the electronic ballast, since, for example, the occurrence of a rectification effect should not lead to the system being switched off during the preheating phase.

According to a preferred embodiment of the present invention, the rectification effect detection takes place in particular in that clock pulses of a (high-frequency) reference clock are counted and compared with one another during the individual half-waves of the lamp voltage or the quantity dependent thereon, the clock pulses counted as a function of the duration of the respective half-wave are. If there is no rectification effect, the clock pulses counted during the positive and negative half-waves match. In contrast, if there is a rectification effect, the clock pulses counted during the positive and negative half-waves differ from one another.

11a shows a circuit-technical implementation of this exemplary embodiment with an up / down counter 107, which receives a signal UZERO as the actual input signal and furthermore as control signals a high-frequency reference clock signal CLK, for example with the frequency 10 MHz, and a reset or reset signal . The signal UZERO assumes a positive and otherwise a negative voltage level during each positive half-wave of the lamp voltage present at the connection VLI and thus detects the zero crossing of the lamp voltage. The counter 107 is designed in particular as a 9-bit counter and is initialized to a medium counter reading, for example to the output counter value N ₀ = 255, when the reset signal is present. The counter 107 is started at the zero crossing of the lamp voltage and counts either up or down during the subsequent half-wave of the lamp voltage. When the measurement signal, ie the lamp voltage, reaches zero again after a half cycle, the counting direction of counter 107 is reversed. After a full period of the lamp voltage has elapsed, the current counter reading N of the counter 103 is connected to a comparator, which can be formed, for example, by the comparator 103 already described above. This comparator 103 compares the current counter reading N with the initialization value or the original counter reading of the counter 107. If there is no rectification effect, it must the counter reading N has reached the initial value N ₀ again after reaching the next zero crossing of the lamp voltage. If, on the other hand, the counter reading N deviates from the initial value N ₀ , there is a rectifying effect. The comparator 103 advantageously compares the counter reading N with the initial value N ₀ within certain tolerance limits, in order not to prematurely conclude that there is a rectification effect. The output signal of the comparator 103 is fed via a D-flip-flop 108 clocked by a latch signal to the measuring phase controller 900, which - as has been described above - evaluates this signal and in particular carries out an event-filtered evaluation, ie only then concludes that there is a rectification effect , if, for example, a rectification effect is reported by the comparator 103 32 times in succession every 255th period of the lamp voltage.

The zero crossing signal UZERO can originate, for example, from a further comparator 203, which monitors the voltage measuring signal present at the voltage connection VLI with regard to its zero crossing. With the aid of this zero voltage comparator 203, the entire integrated measuring system of the control circuit IC2 is cyclically synchronized with respect to the zero point of the lamp voltage. The synchronization advantageously takes place every second period of the output frequency. An exception to this principle is the rectification effect evaluation. In this case, the synchronization is delayed by two further periods over a full period of the lamp voltage due to the integration carried out for the rectification effect evaluation. The output signal of the zero crossing comparator 203 is also fed to the measuring phase control 900 and is of central importance for the control of all controllable switches of the entire control circuit, the actuation of which is controlled in each case to the zero crossing of the lamp voltage.

FIG. 11 b shows a representation of the signal profiles in the circuit shown in FIG. 11 a in the absence of a rectification effect and the states that occur in the process. 1 lb in particular shows that the zero crossing signal UZERO _assumes the positive level during the positive half-wave of the lamp voltage U _VL1 and the counter 107 decreases its counter reading N based on the initialization value NO according to the reference clock CLK until the lamp voltage U _{VL1 passes} through a new zero is present. The counter reading N is then increased again. After a period of the lamp voltage U _vu , the latch signal outputs the output value of the comparator 103 to the measuring phase controller 900 via the D flip-flop 108, and then the counter 107 is reset to the initial value N ₀ using the reset signal. In the case shown in FIG. 11b, after a full period, the _Lamp voltage U _VL1 the counter reading N of the counter 107 again the initial value N ₀ , so that the comparator 103 reports no rectification effect.

11c and 11d, on the other hand, show courses of the counter reading N if there is a rectifying _effect, the counter reading N being greater than N ₀ according to FIG. 11c or smaller than N ₀ according to FIG. 11d after a full period of the lamp voltage U _VL1 and thus the comparator 103 recognizes and reports the rectification effect by comparing N with N ₀ .

As has already been explained above, the comparison of the comparator N advantageously takes place within predetermined tolerance limits, which are defined by threshold values N _S1 and N _S2 according to FIG. 11D, ie the comparator 103 only outputs an output signal corresponding to the rectification effect if the following condition is not fulfilled: N _S2 <N <N _S1 .

The threshold values are advantageously selected asymmetrically in such a way that the distance between N _S1 and N _{0 is} greater than the distance between N ₀ and N _S2 (in particular twice as large), since the control behavior of the electronic ballast occurs when the rectification effect shown in FIG always tries to compensate for the associated decrease in current by changing the frequency. In order to take this behavior into account, the sensitivity for the rectification effect detection at counter readings N which are below the output value N ₀ after a full period of the lamp voltage U _{vu is} increased and the threshold value N _S2 is shifted closer to the output value N ₀ .

A further function block for overvoltage detection of the lamp voltage can be connected to the voltage connection VLI (cf. the arrow shown in FIG. 6), the output signal of this function block also being able to be fed to the measurement phase control 900 and, for example, again event-filtered (cf. the rectification effect evaluation explained above). leads to a corresponding error message to the sequencer 800.

The voltage detection block 200 shown in FIG. 6 comprises a further function block which is provided for the detection of a lamp change. This functional block comprises a sampling circuit 201, a switch S206 and a comparator 202. This lamp change detection circuit makes it possible to detect a change in both the upper gas discharge lamp Gl shown in FIG. 1 and the lower gas discharge lamp G2. So far it was due to circuitry Difficulties only possible and known to monitor and recognize a change in this lower gas discharge lamp G2 by monitoring the lower lamp filament of the lower gas discharge lamp G2. As soon as a change in the lower gas discharge lamp G2 was detected, the entire system was restarted. However, since a change in the upper gas discharge lamp Gl could not be detected, the replacement of this upper gas discharge lamp Gl had no immediate effect, ie no restart was carried out. A fitter was therefore difficult to see which of the two gas discharge lamps Gl, G2 shown in Fig. 1 was actually defective, since even after replacing a defective gas discharge lamp Gl, no automatic restart was carried out, so that the fitter received no feedback as to whether the him replaced gas discharge lamp Gl was actually defective.

With the aid of the present lamp change detection circuit, it is now possible to detect the change of any gas discharge lamp Gl, G2 connected to the electronic ballast. As soon as a lamp change has been recognized, this is communicated via the measuring phase controller 900 shown in FIG. 3 to the sequence controller 800 also shown schematically in FIG. 3, so that it can automatically restart the system after notification of a lamp change. A lamp change is particularly considered if the control circuitry has a lamp fault, e.g. a gas defect that has been identified and reported. In this case, the installer will try to replace the faulty lamp. Initially, however, the fitter does not know which of the gas discharge lamps G1, G2 connected to the electronic ballast is faulty. He will therefore replace one of these connected gas discharge lamps. As soon as the

If the lamp change detection circuit of the voltage detection block 200 shown in FIG. 6 has recognized a lamp change, the sequence control 800 shown in FIG. 3 will restart the system. If a lamp fault is still detected or all the connected gas discharge lamps cannot be ignited, the control circuit switches back to an error or lamp change detection state without the connected gas discharge lamps being able to be operated continuously. For the fitter, this means that the gas discharge lamp that he replaced was either not defective or that another defective gas discharge lamp exists. In this case, the fitter has to replace another gas discharge lamp connected to the electronic ballast. If a successful restart of the system is possible after changing the lamp, this means for the fitter that on the one hand the gas discharge lamp that he replaced was faulty and on the other hand that all are now connected to the electronic ballast Gas discharge lamps are faultless. Overall, the fault detection and troubleshooting for the fitter is significantly simplified in this way, since the fitter can decide immediately after changing a gas discharge lamp based on a successful or unsuccessful restart of the system whether all lamps connected to the system are now faultless or not.

With the aid of the lamp change detection circuit shown in FIG. 6, a lamp change is recognized in that a supply voltage of a certain frequency is applied to the load circuit by the inverter and the transient response of the load circuit is evaluated in this regard. The transient response of the load circuit is in turn assessed on the basis of the measurement signal present at the voltage connection VLI and proportional to the lamp voltage, this measurement signal being sampled several times and thus the characteristic curve of the lamp voltage resulting from the applied supply voltage being assessed.

The supply voltage applied to the load circuit in lamp change detection mode has, in particular, a relatively low frequency of, for example, 40 Hz. Furthermore, only one of the two inverter switches T2, T3 (cf. FIG. 1) is switched on or off alternately with the aforementioned frequency in the lamp change detection mode, while the other inverter switch remains permanently open during the lamp change mode. In the present exemplary embodiment, it is the upper inverter switch T2 that is permanently open, while the lower inverter switch T3 is alternately switched on and off with the low repetition frequency of approximately 40 Hz.

The function of the lamp change detection circuit shown in Fig. 6 is as follows.

As has already been explained, when a fault is detected, the lower inverter switch T3 of the inverter D shown in FIG. 1 is switched on and off with a low repetition frequency of approximately 40 Hz, while the upper inverter switch T2 remains permanently switched off. Because the inverter switch T3 is switched on and off, there is a certain transient response in the load circuit of the electronic ballast, which depends in particular on the gas discharge lamps connected to the electronic ballast. This transient response of the load circuit is reflected in the measurement signal detected via the input connection VLI, which is evaluated by the lamp change detection circuit. For this purpose, the sampling circuit 201 stores certain ones Time T _r T ₃ the current voltage value of the measurement signal present at the connection VLI. In principle, the third measurement at time T _{3 is} not absolutely necessary, but it does increase the reliability of the measurement against interference. The measurement process described above takes place after the inverter switch T3 has been opened and before it has been closed again.

After recording the measuring points at the times T _] - T ₃ , the result is temporarily stored in the downstream digital part (not shown in FIG. 6). The lamp change detection circuit is then reinitialized, ie a specific reference voltage Vrefl 1 is switched on via switch S206 and a new sample value of the voltage signal at terminal VLI is buffered in the sampling circuit 201. The comparator 202 thus carries out a double relative evaluation of the sample values stored in the sampling circuit 201, that is to say the difference between the sample value stored at time T, and the sample value stored at time T ₂ , and the difference between the one recorded at time T j Sample value and the sample value stored at time T ₃ . This evaluation of the relative relationships between the individual sample values is advantageous compared to the evaluation of absolute measured variables, since additional components would be required to evaluate absolute measured variables.

FIG. 7a shows a time diagram of the course of the voltage U _{vu present} at the connection VLI, the switching state of the inverter switch T3 and the switching state of the switch S206 shown in FIG. 6. Furthermore, the individual sampling times T ,, T ₂ and T _{3 are} indicated in FIG. 7a.

The evaluation of the comparison result provided by the comparator 202 between the samples at times T, and T ₂ or Tj and T ₃ takes place in the measuring phase controller 900. On the basis of the transient process, that is to say on the basis of the sample values at times Tj - T ₃ formed voltage characteristic, it can be decided whether one of the gas discharge lamps has been removed during the lamp change detection operation and, if so, which of the gas discharge lamps has been removed. Furthermore, it can be assessed whether instead all lamp filaments of the individual gas discharge lamps are correctly connected to the load circuit, ie all lamps are connected without errors. 7b shows an example of the characteristic curve of the voltage signal U _{VL present} at the connection VLI for three different cases. The characteristic curve a corresponds to the characteristic curve which arises when the upper gas discharge lamp Gl shown in FIG. 1 changes. The characteristic curve b corresponds to the characteristic curve when changing the lower one Gas discharge lamp G2 during lamp change detection operation. The third characteristic curve c shown in FIG. 7b corresponds to the characteristic curve in normal operation without a lamp change, ie in the event that all lamps are connected. By evaluating or _acquiring the characteristic of the voltage signal U _{VL1 present} at the connection VLI, the control _circuit can thus assess which of the connected gas discharge lamps Gl, G2 has been replaced. This means that not only a change in the lower gas discharge lamp G2, but also a change in the upper gas discharge lamp Gl can be reliably detected. As soon as the control circuit has recognized a change in one of the gas discharge lamps connected to the electronic ballast, the system is automatically restarted in order to ignite the connected gas discharge lamps.

In practice, the control circuit IC2 will monitor the transient behavior with regard to the occurrence of the characteristic curves a or b when a lamp fault occurs in an error state. As soon as the voltage at the connection VLI runs in accordance with one of these characteristics, this means that one of the connected gas discharge lamps has been removed from its version for troubleshooting. The control circuit IC2 or sequence control 800 then changes to the actual lamp change detection state, in which, as in the fault state, only the lower inverter switch T3 is opened and closed, for example at 40 Hz, while the upper inverter switch T2 is permanently open. In this state, the control circuit IC2 waits for the appearance of the characteristic curve c, d. H. that a replacement lamp has been used instead of the removed lamp and now all lamps are reconnected. The system then restarts or restarts. This process will be explained again later with reference to FIG. 9.

8a and 8b show two variants of the circuit 300 shown in FIG. 3 for detecting a warm / cold start operation. Both variants have in common that the voltage potential present at the connection NP of the control circuit is always evaluated and it is determined by comparison with a predetermined reference voltage Vrefl 2 whether a warm or cold start is to be carried out. This comparison is carried out with the aid of a comparator 301, the positive measuring input of which is connected to the connection NP. On the output side, the comparator 301 is connected to a state hold circuit 302, which can be implemented, for example, by a D flip-flop. This state hold circuit 302 has the effect that the output signal of the comparator 301 is only switched through and evaluated to the sequence control 800 if a corresponding enable signal EN is present. This enable signal EN takes only at a high level for a short time only when the entire system is restarted or restarted, for example by actuating an appropriate power switch. At no later point in time does a signal change at the NP port result in a state change at the output port of the state hold circuit 302.

In the variant shown in FIG. 8a, by connecting a series resistor R _v either to the high supply voltage potential VDD or to the ground potential, it is possible to switch between a cold start and a warm start operation. If the series resistor R _{v is connected} to VDD, a cold start operation is activated, ie the connected gas discharge lamps are ignited without preheating operation. On the other hand, if the series resistor R _{v is connected} to the ground potential, a warm start operation is carried out, ie the connected gas discharge lamps are ignited with an upstream preheating operation for preheating the lamp electrodes. The comparator 301 can determine whether the resistor R _{v is connected} to the supply voltage potential VDD or the ground potential by monitoring the voltage potential at the connection NP. The evaluation of the comparator output signal finally takes place in the sequence control 800 shown in FIG. 3, which, depending on whether a cold start or warm start mode is selected, controls the gas discharge lamps without or with preheating operating states.

8b shows a variant of the circuit explained above, which enables a dynamic switchover between a warm and cold start operation. The circuit shown in Fig. 8b corresponds essentially to the circuit shown in Fig. 8a, with the exception, however, that a switch S301 is provided internally at the input terminal NP, via which the supply voltage potential VDD can be applied to the input terminal NP, while externally an RC element consisting of the resistor R22 and capacitor C17 already shown in FIGS. 1 and 2 is connected to the connection NP. As in the variant shown in FIG. 8a, the voltage potential present at the input connection NP is monitored by the comparator 301. The operation of the circuit shown in Fig. 8b is as follows.

During normal operation of the electronic ballast, the switch S301 is closed so that the capacitor C17 is charged by the supply voltage potential VDD applied to the input terminal NP. If the system is switched off (for example as a result of a fault) or the system supply is switched from mains to emergency power operation, switch S301 is opened and capacitor C17 discharges with the time constant defined by the RC element. According to the standard, the RC element is advantageously designed in such a way that the capacitor C17 carries the charge can hold until the voltage applied to the input connection NP is greater than the reference voltage Vrefl 2 applied to the comparator 301 for a period of up to 400 ms.

When the system is restarted or restarted, the enable signal EN of the state hold circuit 302 assumes a high level, so that the comparison result of the comparator 301 is switched through. If, at this point in time, the voltage potential present at the input connection NP is still greater than the reference voltage Vrefl2, the sequence control 800 ensures the start-up of the connected gas discharge lamps without preheating operation and thus carries out a cold start. If, on the other hand, the voltage potential present at the input connection NP is less than the reference potential Vrefl2, the connected gas discharge lamps are preheated and a warm start is thus carried out.

From the above description it can be seen that the voltage potential present at the input terminal NP of the control circuit depends on the on-time of the switch S301, which is equivalent to the operating time of the electronic ballast. This variable is decisive for the state of charge of the capacitor C17. Furthermore, the voltage potential at the input connection NP depends on the switch-off time of the switch S301 or the duration of the emergency power operation of the electronic ballast and the time constant of the RC element. These variables are decisive for the discharge process of the capacitor C17.

The circuit shown in FIG. 8b therefore performs a cold or warm start depending on the duration of the switch-off time and on the time constant of the RC element.

By appropriately dimensioning the time constant of the RC element, the one

Switch-off duration can be specified, which is just sufficient for a cold start operation of the connected lamps. For this purpose, the RC element only has to be dimensioned such that, after charging the capacitor C17 and opening the switch S301, the voltage potential applied to the input terminal NP is greater than the reference potential just after the aforementioned switch-off period

Vrefl 2 of the comparator 301 is. However, the maximum permissible time between switching to emergency power operation and restarting or restarting the electronic ballast without preheating the lamp electrodes is set to 400 ms. Accordingly, the resistor R22 and the capacitor C17 are to be dimensioned in such a way that the aforementioned period of 400 ms can be observed. Of course, instead of the RC element shown in FIG. 8b with the resistor R22 and the capacitor C17, any other energy storage circuit can be used, which stores energy depending on the supply voltage potential present at the input terminal NP and discharges with a certain time constant after the supply voltage potential has been disconnected . This energy storage circuit can thus contain any delay elements, as long as there is a defined and known temporal behavior of the delay element or the energy storage circuit.

The function blocks 400 and 500 shown in FIG. 3 will be explained in more detail below. The voltage regulator function block 400 generates an internally regulated, very precise supply voltage VDD for all internal function blocks, which at the same time represents the source for all required reference voltages. As can be seen from FIGS. 1 and 2, this internal supply voltage VDD is applied to the outside via the connection VDD and filtered via the external capacitor C7 with good high-frequency properties. Due to the provision of the internal supply voltage VDD, a single low-voltage level can be used for all functional parts of the entire electronic ballast, which is particularly advantageous for cost reasons.

The reference voltage generator 500 serves for the central generation of all reference quantities for the control circuit IC2, i.e. to generate all reference potentials and reference currents.

The oscillator 600 shown in FIG. 3 represents the central clock source for the entire control circuit IC2. The oscillator 600 is constructed in such a way that no external components are required. The basic clock of the oscillator is set with the help of micro fuses to the desired value of, for example, 10 MHz with an accuracy of z. B. 4-bit matched. The frequency of the clock generator can be reduced to approximately 1/20 of the nominal clock rate, ie to approximately 550 kHz, via a digital input of the oscillator 600. This reduced clock rate, as will be explained in more detail below, is required for certain operating states, in particular for the fault and lamp change detection state, in which the supply energy has to be reduced. Depending on the basic clock of the oscillator 600, the time base generator 700 likewise shown in FIG. 3 generates a plurality of constant time intervals which are supplied to the individual function blocks of the control circuit IC2 via digital outputs of the time base generator 700. The sequential control function block 800 receives, for example, all the time reference quantities from the time base generator 700. All the time quantities generated by the time base generator 700 are a multiple of this Basic clock of the oscillator 600. The time reference variables generated by the time base generator 700 can include, for example, the individual preheating times or the ignition time. As will be explained in more detail below, these temporal reference variables are particularly important for the temporal operating state control of the control circuit IC2, which is carried out by the sequence control function block 800.

The function of the sequence controller 800 will be explained in more detail below with reference to FIG. 9.

The sequence control function block 800 controls the operation of the electronic ballast, for example in accordance with the state diagram shown in FIG. 9. 9, each possible operating state is illustrated by a circle, while the individual arrows represent possible changes in state which occur when a condition associated with the two operating states is met. These conditions are in each case linked to specific states of certain state or monitoring variables of the electronic ballast or the lamp (s), these monitoring variables being processed internally by the sequence control 800 in the form of variables which depend on whether the monitoring variable assumes the corresponding state or not, for example assumes the value "1" when the assigned state is taken or "0" when the state is not taken. The individual variables monitored by the sequence controller 800 can include, for example, time-based variables or error variables. With regard to the time-based variables, the course of a commissioning time, a preheating time, an ignition time or a delay time for the rectification effect detection can be monitored. With regard to the error quantities, for example, the occurrence of a capacitive current in the load circuit (via the current detection block 100), the presence of an overvoltage at the connected gas discharge lamp, the occurrence of a rectification effect or asymmetrical lamp operation, the absence of a lamp or the occurrence of a synchronization error with regard to the zero crossing the lamp voltage (in each case via the voltage detection block 200) are monitored. Furthermore, the output signal of the function block 300 can be monitored, with the aid of which a distinction can be made between a warm and a cold start operation. Any other monitoring parameters of the electronic ballast are of course also conceivable. As has already been explained above, the individual error quantities are detected by the blocks 100-300 shown in FIG. 3, but processing is first carried out by the measuring phase control function block 900 before the individual error quantities are actually evaluated by the sequence control 800. For this purpose, the measuring phase control contains a digital event filter assigned to the corresponding error size for each monitored error size. In principle, this digital event filter performs the function of a counter which counts the uninterrupted occurrence of the corresponding error. An error message is only passed on to the sequence control 800 by the corresponding event filter when the corresponding error has occurred n times in succession, where n corresponds to the filter depth of the corresponding digital event filter and can be different for each error size. As soon as the error no longer occurs during a query, the counter status of the digital event filter is reset and the counting process starts again from the beginning. In this way it is ensured that the sequential control unit 800 does not react prematurely to the occurrence of a specific error, and an operating state change as a result of a specific error message is only carried out when it can be assumed with a relatively high degree of certainty that the corresponding error actually exists.

A special feature in this regard is the digital event filter for the rectification effect detection, since the rectification effect is a creeping, i.e. error occurs slowly over time. The event filter assigned to the rectification effect is therefore dimensioned in such a way that a rectification effect only occurs and a corresponding error message is output to the sequential control unit 800 if the measuring phase control 900 reports a rectification effect 32 times in succession every 255th period of the lamp voltage. Accordingly, the event filter assigned to the rectification effect comprises a filter depth of n = 32 x 255. On the other hand, a filter depth of 64 can be used for the detection of a capacitive current, a filter depth of 3 for the detection of an overvoltage and one for the detection of a synchronization error and for the lamp change detection Filter depth of 7 can be provided. Of course, other filter depth values are also conceivable.

If the following is spoken of the occurrence of a specific error case, the corresponding error message from the measurement phase control 900 to the sequence control 800 after passing through the correspondingly assigned event filter is meant. The initial state of the operating state control shown in FIG. 9 is the so-called. Reset state (state I). The system is in state I whenever the electronic ballast has been started or restarted, which is synonymous with the occurrence of the enable signal EN explained with reference to FIG. 8. To this end, the sequencer 800 may include a hysteresis comparator that monitors the external supply voltage signal VCC within certain limits and generates the enable signal EN if the supply voltage signal VC is within the required range

Supply voltage range is. In this way, the comparator also monitors the switching on and off of the entire system. The enable signal EN can thus occur asynchronously to all other signals depending on the switching on and off of the overall system, with after the appearance of the enable signal EN, i.e. after inclusion The electronic ballast is switched on again and the individual function blocks of the control circuit IC2 are compared. This comparison is carried out by reading in the respective values for the individual micro fuses. These micro fuses are small fuses that are used, for example, to balance the individual internal power sources. Furthermore, as has been explained with reference to FIG. 8, when the enable signal EN occurs, the output signal of the function block 300 shown in FIG. 3 is read in, so that it is determined at this point in time whether the connected gas discharge lamps with a cold or warm start in To be put into operation. Overall, in state I the control circuit IC2 is thus initialized.

After the control circuit has been initialized, the sequential control system 800 automatically changes to a commissioning state (state II). Exceptionally, the transition from state I to state II is not linked to certain conditions and occurs automatically each time the electronic ballast is restarted or restarted. In state II, the harmonic filter starts up or the load circuit of the electronic ballast settles. Furthermore, the coupling capacitor of the load circuit is precharged in state II. In this phase, all error detectors are deactivated, i.e. there is no evaluation of the aforementioned error sizes.

A preheating state III is started from state II, for example if a start-up time assigned to state II, which denotes the normal operating time of state II, has expired and no cold start operation has been reported by function block 300 shown in FIG. 3. If, on the other hand, the start-up time has not yet expired, the system remains in state II, which is shown in FIG. 9 by a state which starts from state II and returns to state II Arrow is shown. If a cold start operation was detected by the function block 300 and the start-up time has already expired, the sequence control 800 changes directly from the state II to an ignition state IV, which corresponds to the previously explained warm start operation.

In the first preheating state III, the inverter half bridge is driven in such a way that it oscillates at the upper limit in terms of frequency and generates, for example, an output frequency of approximately 80 kHz. In this state, the preheating control, the overvoltage detection and the capacitive current detection can be activated.

After a predefined preheating time has elapsed, a switch is made to the ignition state IV already mentioned, if no lamp overvoltage and no capacitive current operation have been detected. During the ignition state IV, all error detectors of the control circuit with the associated event filters of the measurement phase control function block are deactivated. Accordingly, starting from this state IV, it is also not possible to jump to a fault state VII, which will be explained in more detail below. This means that the system remains in ignition state IV until the state change condition for the transition to an operating state V is fulfilled.

In the ignition state IV, the working frequency of the inverter of the electronic ballast can be changed depending on the value of the lamp current detected and the states of the overvoltage and capacitive current detection. Starting from the operating point of the load circuit specified by the preheating mode, the control variable "lamp current" is used to try to reduce the output frequency of the inverter because the detected lamp current is significantly too small compared to the specified setpoint due to the fact that the ignition has not yet taken place. This control process becomes until the overvoltage detection or capacitive current detection prevents or counteracts the continuous reduction of the inverter frequency. As a rule, the overvoltage detection will become dominant as an influencing factor. As a result of the dominance of the overvoltage detection that is now present, the lamp voltage is regulated, as it were. This behavior changes until the lamp is ignited or until the specified ignition time has elapsed, but as a rule the gas discharge lamp will ignite before the specified ignition time has elapsed, in which case the lamp current surge becomes dominant again and the output frequency of the inverter is reduced until the stable operating point specified by the lamp current reference value has been reached. The capacitive current detection in ignition state IV is only in the event of a fault, for example when saturation of the resonant circuit choke L3 shown in FIG. 1, actively intervene in the ignition process. As soon as the capacitive current detection responds, the output frequency of the inverter is pushed up by the control circuit until another of the aforementioned influencing variables becomes dominant again during the ignition operation IV. In addition, it should be pointed out at this point that during the ignition state IV of the control circuit of the control circuit IC2, for example every eighth period of the output frequency of the inverter, a new setpoint / actual value comparison is carried out, since it has been found in practice that with the help of such a reduced controller cycle, for example, the lamp voltage can be corrected with a significantly lower ripple.

The ignition state IV can only be left in the direction of the previously mentioned operating state V after the specified ignition time has elapsed. This change of state is in particular independent of whether control in the ignition state IV is still based on the ignition voltage or is already based on the lamp current.

After reaching the operating state V shown in Fig. 9, as already explained, the mean or effective value of the lamp current is regulated, i.e. the output frequency of the inverter depends on the detected lamp current. Overvoltage, capacitive current and synchronization error detection is activated during this operating state V, the control circuit also carrying out a new setpoint / actual value comparison only every second period of the inverter output frequency during this state. The rectification effect detection (GLRE) can also be activated. The operating state V is not limited in time, i.e. is in principle an endless loop and can only be exited when one of the activated error detectors responds. All error detectors of the control circuit are advantageously activated during the operating state V.

If an error now occurs in one of the previously explained states III or V, which has been detected by the corresponding state-dependent activated fault detector, the fault state VII shown in FIG. 9 is started up. This fault condition VII is therefore the central point of contact for all serious operational disturbances. Fault state VII is jumped on directly from preheating state III if an overvoltage or a capacitive current operation has been detected during these preheating states. In contrast, fault state VII is started from operating state V if, during this state, capacitive current operation, an overvoltage fault, a synchronization fault and / or the occurrence of a Rectification effect, etc. has been detected with respect to the connected gas discharge lamps.

The start-up of the fault state VII can, for example, be associated with a corresponding signaling of the respective fault for the user. Fault status VII is only exited by the sequential control system if, after restarting the system, start-up status II is restarted via reset status I and the gas discharge lamps are started up again. Alternatively, error state VII can be left if it is detected in this state that not all of the lamps connected to the electronic ballast have intact lamp filaments. This is equivalent to leaving the fault state VII in the direction of the lamp change detection state VIII already mentioned as soon as one of the connected gas discharge lamps is removed from its socket. In addition, it should be pointed out that the operating current consumption of the control circuit is reduced to a minimum possible value during the fault state VII.

In the fault state, the electronic ballast is operated as in the lamp change detection state, i. H. the lower inverter switch T3 is opened and closed at a low frequency of, for example, 40 Hz, while the upper inverter switch is permanently opened. As has already been explained with reference to FIG. 6/7, the control circuit IC2 waits in the fault state VII for the occurrence of the voltage characteristic curves a or b (cf. FIG. 7a) at the voltage measurement connection VLI, which corresponds to the removal of one of the connected gas discharge lamps Gl, G2 . In this case, the control circuit IC2 changes to the lamp change detection state VIII.

With the help of the lamp change detection method already explained above, the control circuit can reliably detect both a change or a removal of the upper gas discharge lamp Gl and the lower gas discharge lamp G2 (cf. FIG. 1) and automatically restart the system after detecting a lamp change. While it is checked in the fault state VII whether one of the gas discharge lamps has been removed, the lamp change detection state VIII monitors whether all the gas discharge lamps have been inserted. As soon as it has been recognized that all gas discharge lamps have been inserted, ie all lamp filaments connected to the electronic ballast are intact, the system automatically returns to commissioning state II switched over and the gas discharge lamps started again according to the functional circuit shown in FIG. 9. All other fault detectors are also deactivated during lamp change detection state VIII, with the exception of lamp change detection.

Finally, the function of the inverter control 1000 shown in FIG. 3 will be briefly explained below.

The inverter control function block 1000 is used to generate control signals for the upper and lower inverter switches T2, T3 (cf.

1), which are output via the output connections OUTH or OUTL of the control circuit. Depending on these control signals, the two inverter switches are either switched on or opened. As a rule, the inverter control generates

1000 alternating control pulses for the control connections OUTH and OUTL of the two inverter switches T2 and T3 and can also be an internal one

Have a dead time counter function to ensure a sufficient dead time between the activation of the two inverter switches. In the lamp change detection state

VIII (see FIG. 9), the inverter control 1000 ensures that the upper

Output terminal OUTH, the upper inverter switch T2 remains permanently open, while only the lower inverter switch T3 with a relatively low one

Frequency is opened and closed alternately via the lower output connection OUTL.

The inverter control 1000 in particular ensures an asymmetrical duty cycle of the inverter switches, but this asymmetry is only 2.1% for an output frequency of the inverter of 43 kHz, for example, and only 4% for an output frequency of 80 kHz, and is therefore hardly significant. The generation of asymmetrical output signals for the two inverter switches leads to an increase in the frequency resolution of the inverter, i.e. With the help of the control circuit, smaller frequency steps of the inverter can be set.

The generation of an asymmetrical duty cycle also has the effect that the so-called. "Walmen" of the connected gas discharge lamps can be changed. This Walmen is an effect of "running layers", which occurs especially at low temperatures shortly after the start of the system, which is due to an uneven light distribution in the corresponding gas discharge lamp. These "running layers" consist of light / Dark zones that run along the lamp tube at a certain speed. As is known, for example, from EP-B 1-0490 329, this running effect can be soiled by superimposing a small direct current in such a way that it no longer has a disruptive effect. The generation of an asymmetrical duty cycle by the present control circuit of the electronic ballast can also prevent the occurrence of the so-called. Counteract "Walmens".

As has already been explained above, with the aid of the present control circuit, an asymmetrical duty cycle for the two inverter switches is generated during individual half-periods, although the duty cycle is averaged over an entire period. Since only asymmetrical output signals are to be generated in the operating state V shown in FIG. 9, the inverter control 1000 evaluates, for example, a corresponding control signal which only releases the asymmetrical operation (for example by assuming a high level) if the system is in the operating state V is located.

Claims

Expectations

1. Electronic ballast for operating at least one gas discharge lamp, with an AC voltage source (D), with a load circuit (E) driven by the AC voltage source (D), which contains at least one gas discharge lamp (Gl, G2), and with a control circuit (IC2) for Controlling the operation of the gas discharge lamp (Gl, G2), characterized in that the control circuit (IC2) monitors a circuit size of the electronic ballast and, depending on the value of this circuit size when starting the electronic ballast, the gas discharge lamp (Gl, G2) either with a preheating mode Preheating the lamp filaments of the gas discharge lamp (Gl, G2) or without preheating.

2. Electronic ballast according to claim 1, characterized in that the circuit size of the electronic ballast is at a terminal (NP) of the control circuit (IC2) applied voltage potential.

3. Electronic ballast according to claim 2, characterized in that the control circuit (IC2) comparator means (301) for comparing the voltage potential applied to the control circuit (IC2) with a reference value

(Vrefl2).

4. Electronic ballast according to claim 3, characterized in that the control circuit (IC2) operates the gas discharge lamp without the preheating operation if the comparator means (301) indicate a voltage potential applied to the control circuit (IC2) which is greater than the reference value (Vrefl2) , and that the control circuit (IC2) in the other case operates the gas discharge lamp (Gl, G2) with the preheating mode.

5. Electronic ballast according to claim 4, characterized in that the reference value applied to the comparator means (301) is approximately IV.

6. Electronic ballast according to one of claims 3-5, characterized in that the output signal of the comparator means (301) of a state hold circuit

(302) is supplied, which only switches the output signal of the comparator means (301) through to its output when the electronic ballast is started, the control circuit (IC2) optionally also using the output signal of the state hold circuit (302) for starting up the gas discharge lamp (Gl, G2) or evaluates without preheating.

7. Electronic ballast according to claim 6, characterized in that the state hold circuit (302) comprises a D flip-flop, at the

A clock signal is present which only temporarily has a high level when the electronic ballast is started.

8. Electronic ballast according to one of the preceding claims, characterized in that the circuit size monitored by the control circuit (IC2) is a selectable supply voltage potential (VDD) via a resistance circuit (R _v ) to the control circuit (IC2).

9. Electronic ballast according to one of the preceding claims, characterized in that the control circuit (IC) monitors the state of charge of an energy storage circuit (R22, C17) as the monitored circuit size of the electronic ballast, which is charged during operation of the gas discharge lamp (Gl, G2) and discharges when the electronic ballast is switched off according to a certain time constant.

10. Electronic ballast according to claim 9, characterized in that the energy storage circuit one to a connection (NP) of the control circuit

(IC2) connected series circuit with a charging capacitor (C17) and a resistor (R22).

11. Electronic ballast according to claim 9 or 10 and one of claims 3-7, characterized in that the energy storage circuit (R22, C17) has such a time constant that that monitored by the Steuererschalmng (IC2) and the state of charge of the

Energy storage circuit-dependent voltage potential after switching off the electronic ballast for a period of approx. 400 ms is greater than the reference value (Vrefl 2) applied to the control shutter (IC2).

12. Electronic ballast according to one of claims 9-11, characterized in that the energy storage circuit (R22, C17) is connected via a controllable switch (S301) to a supply voltage source (VDD), the energy storage circuit (R22, C17) by closing the controllable switch (S301) is charged.

13. Electronic ballast according to claim 12, characterized in that the control circuit (IC2) controls the controllable switch (S301) in such a way that when the electronic ballast is started, the switch is first opened and only closed after the gas discharge lamp (Gl, G2) has been ignited and again is opened when the electronic ballast is switched off.