Publication number | US20030137359 A1 |

Publication type | Application |

Application number | US 10/053,250 |

Publication date | 24 Jul 2003 |

Filing date | 18 Jan 2002 |

Priority date | 18 Jan 2002 |

Also published as | CN1280991C, CN1433152A, EP1330035A2, EP1330035A3, US6600378 |

Publication number | 053250, 10053250, US 2003/0137359 A1, US 2003/137359 A1, US 20030137359 A1, US 20030137359A1, US 2003137359 A1, US 2003137359A1, US-A1-20030137359, US-A1-2003137359, US2003/0137359A1, US2003/137359A1, US20030137359 A1, US20030137359A1, US2003137359 A1, US2003137359A1 |

Inventors | Jari Patana |

Original Assignee | Nokia Corporation |

Export Citation | BiBTeX, EndNote, RefMan |

Referenced by (7), Classifications (6), Legal Events (6) | |

External Links: USPTO, USPTO Assignment, Espacenet | |

US 20030137359 A1

Abstract

A fractional-N frequency synthesizer is disclosed wherein the multi-modulus frequency divider in the feedback path of the phase locked loop is controlled by a delta-sigma modulator to achieve the desired division ratio. The fractional input control signal to the delta sigma modulator is dithered to break any periodicity in the modulator output signal to avoid the generation of fractional spurious frequencies.

Claims(20)

a phase locked loop including a phase frequency detector, a loop filter, a voltage-controlled oscillator and a multi-modulus frequency divider in a feedback loop between the voltage-controlled oscillator output and an input of the phase frequency detector;

a delta-sigma modulator having an input for receiving a fractional control word and an output coupled to the multi-modulus frequency divider for controlling the division ratio of the multi-modulus frequency divider in response to the input fractional control word;

a generator for producing a signal in accordance with and related to a frequency compensation loop error signal from the multi-modulus frequency divider; and

means for adding said generator signal output to a fractional input control word to produce a zero average dither fractional control word as the input to the delta-sigma modulator to generate a multi-modulus input control signal whereby the division ratio is changed without generation of fractional spurious frequencies.

said fractional input control word further comprises a separate F_{input }control signal and an M_{input }control signal;

means for combining said F_{input }control signal and said generator signal output to produce a delta-sigma modulator input control word, said delta-sigma modulator generating an output control word in response to said input control word; and

means for combining said M_{input }control signal and said delta-sigma modulator output control word to generate said multi-modulus frequency divider division ratio control.

said fractional input control word further comprises a separate F_{input }control signal and an A_{input }control signal and an N_{input }control signal;

means for combining said F_{input }control signal and said generator signal output to produce a delta-sigma modulator input control word, said delta-sigma modulator generating an output control word in response to said input control word;

means for combining said A_{input }control signal and said delta-sigma modulator output control word to generate a multi-modulus “A” control input signal; and

means for combining said N_{input }control signal and the result of the combined A_{input }control signal and delta-sigma modulator output control word to generate a multi-modulus “N” control input signal.

providing a fractional-N frequency synthesizer comprising a phase locked loop including a phase frequency detector, a loop filter, a voltage-controlled oscillator and a multi-modulus frequency divider in a feedback loop between the voltage-controlled oscillator output and an input of the phase frequency detector;

providing a delta-sigma modulator having an input for receiving a fractional control word and an output coupled to the multi-modulus frequency divider for controlling the division ratio of the multi-modulus frequency divider in response to the input fractional control word;

producing a signal in accordance with and related to a frequency compensation loop error signal from the multi-modulus frequency divider; and

adding said signal to a fractional input control word to produce a zero average dither fractional control word as the input to the delta-sigma modulator to generate a multi-modulus input control signal whereby the division ratio is changed without generation of fractional spurious frequencies.

providing a reference frequency;

providing a phase locked loop voltage controlled oscillator;

generating a desired output frequency by controlling the division ratio of a multi-modulus frequency divider in a feedback path in the phase locked loop;

dithering the fractional input control signal to a delta sigma modulator using a sine wave generator; and

controlling the multi-modulus division ratio with the output signal produced by the delta-sigma modulator.

Description

[0001] The present invention relates generally to delta-sigma modulator based fractional-N phase locked loop frequency synthesizers and deals more particularly with a delta-sigma modulator based fractional-N phase locked loop frequency synthesizer with a sine wave generator to break the periodicity of the delta-sigma modulator output to eliminate the generation of fractional spurious frequencies.

[0002] Digital frequency synthesizers have long been used in communication systems, particularly RF communication systems, to generate RF signals carried over RF channels. In frequency synthesis, it is desirable to achieve the selected frequency output in as little time as possible with any spurious outputs minimized. It is known to create a frequency synthesizer by placing a frequency divider function between the voltage-controlled oscillator (VCO) output and the phase frequency detector (PFD) in a phase-locked loop (PLL), wherein the output is an integer-N multiple of the input reference frequency to the PFD. The spurious outputs in question are usually associated with phase detectors and occur at the phase detector operating frequency, which is generally the same as the channel spacing. Incorporating a fractional-N division function in the PLL helps overcome problems of spurious frequency outputs in an integer PLL by allowing the phase detector to operate at a much higher frequency for the same channel spacing.

[0003] A number of methods that are based upon the concept of integer-N frequency synthesis are known to realize the fractional-N division function and include pulse swallowing, phase interpolation, Wheatly random jittering and delta-sigma modulation to control the multi-modulus, including dual-modulus, frequency dividers to provide the division function. Of the known methods, a delta-sigma modulator realization of a fractional-N frequency synthesizer is desirable and preferable to achieve low phase noise, fast settling time, fine channel resolution and wide tuning bandwidth. The delta-sigma modulator fractional-N phase locked loop frequency synthesizer is based on the concept of division ratio averaging, wherein an integer frequency divider rather than a fractional frequency divider is used, and the division ratio is dynamically switched between two or more values, effectively providing a non-integer number division function. One of the more important advantages of using the delta-sigma modulator to control a multi-modulus is the ability to shape phase noise introduced by the delta-sigma modulator controlled fractional-N division function. A problem generally associated with such a delta-sigma modulator fractional-N frequency synthesizer is the appearance or presence of fractional spurious levels at a fractional offset frequency. The fractional spurious levels may also appear at the fractional offset frequency harmonics. The fractional spurious levels in delta-sigma modulator based fractional-N frequency synthesizers may originate from several sources including the operation of the delta-sigma modulator itself, coupling between the multi-modulus prescaler or charge pump driving the loop filter and the outside world through power supply feeds or substrates, and the nonlinearity of the charge pump. The fractional spurious frequencies may also originate from the spacing error or timing error of the multi-modulus prescaler.

[0004] It is a general object therefore of the present invention to provide a method and related apparatus to prevent the generation of spurious frequency errors in a delta-sigma based fractional-N frequency synthesizer.

[0005] It is another object of the present invention to break the periodicity of the multi-modulus control output signal of the delta-sigma modulator to eliminate fractional spurious frequencies in the fractional-N frequency synthesizer originating from the operation of the delta-sigma modulator.

[0006] It is a further object of the present invention to provide a delta-sigma based fractional-N phase locked loop frequency synthesizer with a sine wave generator to break the periodicity of the output signal of the delta-sigma modulator to eliminate the production of fractional spurious frequencies.

[0007] Other objects and features of the present invention will become readily apparent from the following written detailed description taken together with the drawings forming a part thereof.

[0008] The invention resides in a fractional-N frequency synthesizer having a delta-sigma modulator control of the division ratio of a multi-modulus frequency divider in the feedback path of the phase locked loop. The output control signal of the delta-sigma modulator is dithered to break the periodicity of the division ratio control signal which occurs when the fractional control input words to the delta-sigma modulation has too few “zeros” or “ones” which cause the generation of fractional spurious frequencies. The invention avoids the generation of the fractional spurious frequencies.

[0009] In a one aspect of the invention, a delta-sigma fractional-N frequency synthesizer comprises a phase locked loop including a phase frequency detector, a loop filter, a voltage-controlled oscillator and a multi-modulus frequency divider in a feedback loop between the voltage-controlled oscillator output and an input of the phase frequency detector. The delta-sigma modulator has an input for receiving a fractional control word and an output coupled to the multi-modulus frequency divider for controlling the division ratio of the multi-modulus frequency divider in response to the input fractional control word. A generator produces a signal in accordance with and related to a frequency compensation loop error signal from the multi-modulus frequency divider. Means are provided for adding the generator signal output to the fractional input control word to produce a zero average dither fractional control word as the input to the delta-sigma modulator. The delta sigma modulator generates a multi-modulus input control signal whereby the division ratio is changed without generation of fractional spurious frequencies.

[0010] Preferably, the order of the delta-sigma modulator has an integer value in the range of Z to X, where Z is an integer value of at least 2 and X has an arbitrary integer value greater than Z.

[0011] Preferably, the generator output signal frequency has a value in the range of F_{comp}/Z to F_{comp}/Y, where Z is an integer value of at least 2 and the maximum value of Y is related to the loop filter and the frequency compensation loop error signal.

[0012] Preferably, the generator signal output is an asymmetrical signal.

[0013] Preferably, the generator signal output is a symmetrical signal.

[0014] Preferably, the generator is a symmetrical sine wave generator.

[0015] Preferably, the generator is an asymmetrical sine wave generator.

[0016] Preferably, F_{comp}/Z is equal to 4.

[0017] Preferably, F_{comp}/Z is equal to 8.

[0018] Preferably, F_{comp}/Z is equal to 16.

[0019] Preferably, F_{comp}/Z has an arbitrary integer value equal to or greater than 1.

[0020] Preferably, the multi-modulus frequency divider is a dual-modulus frequency divider.

[0021] In a further aspect of the invention, the dual-modulus frequency divider includes a prescaler coupled to the output of the voltage-controlled oscillator and includes an N-divider and A-divider coupled to the output of the prescaler. The prescaler has a division ratio control input coupled to the A-divider to switch the division ratio in response to the A-divider completing a predetermined count.

[0022] In another aspect of the invention, the fractional input control word further comprises a separate F_{input }control signal and an M_{input }control signal. Means are provided for combining the F_{input }control signal and the generator signal output to produce a delta-sigma modulator input control word. The delta-sigma modulator generates an output control word in response to the input control word. Means are also provided for combining the M_{input }control signal and the delta-sigma modulator output control word to generate the multi-modulus frequency divider division ratio control.

[0023] Preferably, the multi-modulus frequency divider further comprises a dual multi-modulus frequency divider.

[0024] In a further aspect of the invention, the fractional input control word further comprises a separate F_{input }control signal and an A_{input }control signal and an N_{input }control signal. Means are provided for combining the F_{input }control signal and the generator signal output to produce a delta-sigma modulator input control word. The delta-sigma modulator generates an output control word in response to the input control word. Means are provided for combining the A_{input }control signal and the delta-sigma modulator output control word to generate a multi-modulus “A” control input signal. Means are also provided for combining the N_{input }control signal and the result of the combined A_{input }control signal and delta-sigma modulator output control word to generate a multi-modulus “N” control input signal.

[0025]FIG. 1 is a schematic functional block diagram of a delta-sigma modulator based fractional-N synthesizer.

[0026]FIG. 2 is a schematic functional block diagram of a first embodiment of a delta-sigma fractional-N synthesizer of the present invention.

[0027]FIG. 3 is a schematic functional block diagram of an alternate embodiment of the delta-sigma fractional-N synthesizer of the present invention.

[0028]FIG. 4 is a schematic functional block diagram of a further alternate embodiment of the delta-sigma fractional-N synthesizer of the present invention.

[0029]FIG. 5 is a schematic functional block diagram of a further alternate embodiment of the delta-sigma fractional-N synthesizer of the present invention.

[0030]FIG. 6 is a functional block diagram of a sine wave signal generator using logic.

[0031]FIG. 7 is a functional block diagram of a sine wave signal generator using ROM.

[0032]FIG. 8 is a functional block diagram of a sine wave signal generator using RAM.

[0033] Turning now to the drawings and first considering FIG. 1, a schematic functional block diagram of a representative delta-sigma modulator based fractional-N phase locked loop frequency synthesizer is illustrated therein and generally designated **10**. The delta-sigma fractional-N frequency synthesizer **10** comprises a phase frequency detector (PFD) **16**, a loop filter **22**, and a voltage-controlled oscillator (VCO) **28**. A reference frequency F_{ref }at the input **12** to the frequency synthesizer **10** is applied to the input **14** of the PFD **16**. A multi-modulus frequency divider **34** is located in the feedback loop between the VCO output **30** and an input **38** of the PFD **16**. The output **18** of the PFD **16** is coupled to the input **20** of the loop filter **22**. The loop filter **22** functionally operates as an integrating capacitor. The output **24** of the loop filter **22** is coupled to the VCO input **26**. The VCO **28** generates a frequency signal F_{out }at the VCO output **30** in response to the signal at its input **26**. The frequency F_{out }at the VCO output **30** is coupled to the input **32** of the multi-modulus frequency divider (MMD) **34**. The multi-modulus frequency divider **34** is also coupled to and controlled by a delta-sigma modulator (DSM) designated generally **42**. The control output **44** of the DSM **42** is coupled to the control input **40** of the MMD **34**. The output **36** of the multi-modulus frequency divider **34** is connected to the input **38** of the PFD **16**. The signal F_{comp }at the input **38** to the PFD **16** is representative of the loop phase error, that is, the difference in phase between the frequency F_{out }and the input frequency F_{ref}. The output signal F_{comp }from the MMD **34** is the phase of the VCO output signal F_{out }divided by the fractional divider or multi-modulus divider division factor M (F_{comp}=F_{out}/M), which is controlled by the DSM **42**. The difference in phase between the frequency F_{comp }and frequency F_{ref }is produced in the output **18** of the PFD **16**. In actuality, the output signal **38** of MMD **34** is a clock signal and the PDF **16** measures the difference between the rising edge of the F_{comp }signal and the rising edge of the F_{ref }signal. Also, the phase difference can be produced using the falling clock edge of the F_{comp }and F_{ref }signals. The PFD **16** is commonly shown in the art as two separate function blocks: a phase detector (PD) and a charge pump (CP) and the reader is referred to text books, literature, data sheets and other information readily available for further explanation of the PFD operation. The PFD **16** measures the phase difference and adjusts (advances or retards) the phase of the VCO **28** and thus the frequency F_{out }produced by the VCO. The frequency F_{out }at the VCO output **30** is related to the input reference frequency F_{ref }by a scaling factor as determined by the MMD **34**.

[0034] The frequency F_{out }at the VCO output **30** is a fraction of the input reference frequency F_{ref}. Since the MMD **34** in actuality is not dividing by a fractional division ratio, but rather an integer value, the fractional-N frequency synthesis is achieved through division ratio averaging, that is, the division ratio is dynamically switched between two or more values, effectively causing the divider to divide by a non-integer number. In FIG. 1, the DSM **42** controls the division ratio of the MMD **34** in accordance with information in an N-bit control word F_{ract }on input line **46** coupled to the input **48** of the DSM **42**. For purposes of understanding in FIG. 1, the input control word F_{ract }includes all the necessary information to be provided to the multi-modulus frequency divider including any pre-dividers. Likewise, the multi-modulus frequency divider may take on different forms and implementations and for purposes of explanation in FIG. 1, the multi-modulus frequency divider produces the loop phase error signal F_{comp}. The clock signal of the DSM **42** is not shown in FIG. 1, however it can be F_{ref}, F_{comp }or an even faster clock signal wherein the maximum clock frequency F_{max }is the VCO output frequency F_{out}. As discussed further herein, it is preferable to use the F_{comp }clock signal as the clock signal which insures the output signal of the DSM is correctly synchronized with the MMD. A drawback and disadvantage of the currently known delta-sigma modulator based fractional-N frequency synthesizer **10** is the generation of fractional spurious frequencies that are created by the periodicity of the signal at the output **44** of the DSM **42** when the input fractional control word F_{ract }has too few “zeros” or “ones.” The condition of too few “zeros” or “ones” occurs because all desired RF channels or VCO output frequencies must be selectable. If the desired channel is only one fractional channel different than or away from the integer channel, then the input word/code F_{ract }to the DSM is “00000001” in binary format when the input word width to the DSM is 8 bits. This causes a large least significant bit (LSB) fractional offset spurious and its harmonics. Correspondingly, the input word/code F_{ract }“11111110” also produces a large least significant bit (LSB) fractional offset. Similarly, if only the most significant bit (MSB) is high “10000000”, then the MSB fractional spurious appears and output signal of the DSM does not have a good noise shape. Empirical measurements and observations of the fractional-N synthesizer show that at least 3 or 4 “zeros” or “ones” in the input code F_{ract }results in good noise shaping features, because the DSM produces a sufficient number of different frequency components/terms. Although greatly improved, it still is not possible to avoid all bad channels, that is, it is not possible to avoid the generation of fractional spurious frequencies and as illustrated in the embodiments shown in FIGS. **2** to **5**, a sine wave generator is added to overcome the problem of too few “zeros” or “ones” which cause a periodicity to the DSM output control signal and the generation of spurious frequency signals.

[0035] Turning now to FIG. 2, a schematic functional block diagram of a first embodiment of a delta-sigma modulator fractional-N phase locked loop frequency synthesizer of the invention is illustrated therein and generally designated **100**. The delta-sigma fractional-N frequency synthesizer **100** comprises a phase frequency detector (PFD) **102**, a loop filter **104**, a voltage-controlled oscillator (VCO) **106** and a multi-modulus frequency divider (MMD) **108** in a feedback loop between the VCO output **120** and an input **130** of the PFD **102**. The signal F_{out }at the output **120** of the VCO **106** is coupled to the input **122** of the MMD **108**. The output **128** of the MMD **108** is coupled to the input **130** of the PFD **102**. The signal F_{comp }at the input **130** to the PFD **14** is representative of the loop phase error, that is, the difference in phase between the frequency F_{out }and the input frequency F_{ref }as discussed above. The PFD **102** functions to measure the phase difference between the input frequency F_{ref }and the VCO output frequency F_{out }and adjusts (advances or retards) the phase of the VCO and thus the frequency F_{out }produced by the VCO. The MMD **108** is also coupled to and controlled by a delta-sigma modulator (DSM) designated generally **110**. The control output **124** of the DSM **110** is coupled to the control input **126** of the MMD **108**. The VCO output frequency F_{out }at the VCO output **120** is a fraction of the reference frequency F_{ref }on the line **112** connected to the input of the PFD **102**. The output **118** of the loop filter **104** is coupled to the input of the VCO **106**. The output **116** of the PFD **102** is coupled to the input of the loop filter **104**, which functionally operates as an integrating capacitor.

[0036] The output **128** of the MMD **108** is also coupled to the input **134** of a sine wave generator designated generally **132**. The output **136** of the sine wave generator **132** is coupled to an adder **138**. The fractional N-bit control word F_{ract }on the lead **140** is coupled to the input **146** of the adder **138** wherein the fractional N-bit control word is added to the output signal of the sine wave generator **132** to produce a varying (N+1)-bit control word at the output **142** of the adder **138**.

[0037] The output **142** of the adder **138** is coupled to the input **144** of the DSM **110**. The input control signal to the delta-sigma modulator is “dithered” by adding a symmetric “average zero” signal produced by the sine wave generator to the fractional N-bit control word F_{ract}. The “dithered” input signal to the DSM **110** breaks the periodicity of the signal at the output **124** which is coupled to the input **126** of the MMD **108** to control the division ratio and eliminates the otherwise produced fractional spurious signals. As mentioned above, without dithering, the output of the DSM is too repetitive or periodic in cases where there are too few “zeros” or “ones”. For example, in one fractional-N frequency synthesizer model with a third order DSM and a F_{ract }input word in which only the MSB is “one” and the other remaining bits are “zero”, the DSM produces only 4 different control values (0,2,−1,1), which repeat continually. Thus, the DSM cannot provide sufficient noise shaping without dithering the F_{ract }input signal to the DSM. It also depends on a specific application as to which input bit combination produces the worst fractional spurious due to the periodicity of the DSM output.

[0038] Turning now to FIG. 3, a delta-sigma fractional-N phase locked loop frequency synthesizer embodying the invention is illustrated therein and designated generally **200**. The delta-sigma fractional-N synthesizer **200** comprises a phase frequency detector (PFD) **202**, a loop filter **204**, a voltage-controlled oscillator (VCO) **206** and a multi-modulus frequency divider (MMD) **208** in the feedback loop between the VCO output **210** and an input **212** of the PFD **202**. A reference frequency F_{ref }is coupled to the input **214** of an R-divider designated generally **216**. An R control signal on the R-input **218** causes the R-divider to be loaded with the desired count to scale the frequency of F_{ref}. The function of the R-divider is to scale the reference frequency F_{ref }to a lower frequency and is typically under the control of a digital signal processor (DSP) that provides the R-input signal. The R-divider allows use of the same phase locked loop PLL in different solutions, for example, a given used crystal oscillator may be changed without changing the desired channel spacing. Various different RF specifications or protocols (GSM, PDC, WCDMA) in the same wireless telephone may require a change to the value of the R-divider to achieve the desired frequency. It will be recognized by those skilled in the art that the highest comparison frequency generally provides the best performance of the PLL.

[0039] The output **220** of the R-divider **216** is coupled to one input **222** of the PFD **202**. The frequency F_{out }produced by the VCO **206** is a fraction of the input frequency F_{ref }as input to the PFD **202** from the output **220** of the R-divider **216**. The VCO output **210** is coupled to the input **224** of the MMD **208**. The F_{comp }signal from the output **226** of the MMD **208** is coupled to a clock input **228** of a DSM **230** and a clock signal input **232** of a sine wave generator **234** in addition to the input **212** to the PFD **202**. The output **236** of the sine wave generator **234** is coupled to one input **238** of an adder **240** where the sine wave signal is combined with a fractional-N-bit control word F_{input }on the lead **242** coupled to the input **262** of the adder **240** to produce a “dither” control signal at the output **244** of the adder **240**. The output **244** is coupled to the input **246** of the DSM **230**. The DSM **230** produces a control signal at its output **248** which is coupled to the input **250** of an adder **252**. An M_{input }word on the lead **256** is coupled to the input **254** of the adder **252** where it is combined with the output of the DSM **230** to produce a multi-modulus control word at the adder output **258**. The adder output **258** is coupled to the control input **260** to the MMD **208** to control the division ratio. The fractional input signal F_{ract }on lead **46** in FIG. 1 and on lead **140** in FIG. 2 is divided into two separate control input signals F_{input }and M_{input }in FIG. 3 because the F_{ract }input signal is only coupled to the input of the DSM in FIGS. 1 and 2. In many applications, M is an integer value and F is a fractional value. Finally, the output of the DSM is combined with the M input before the MMD. Note, that in FIG. 3, it is expected that the resultant of adder **240** never overflows the capacity of the input of the DSM and the amplitude of the sine wave generator (bit number) is less than the bit number of the F_{input }signal.

[0040] The sine wave generator **234** produces a fixed frequency at its output **236** and is related to the comparison frequency F_{comp }appearing at the sine wave generator input **232**. The comparison frequency F_{comp }can be divided by a factor in the range of 2 to a value Y. The value of Y depends on the loop filter used and the comparison frequency F_{comp }to determine which sine frequency value is the best choice to achieve the needed performance. Normally, the highest possible frequency is the best choice because the generated frequency will be as far as possible outside the loop. If the sine wave frequency is for example, F_{comp}/2, then there are only two different values in the output of the sine wave generator, 1 and −1 which is not a sine wave signal. At least four values are needed to produce a sine wave. The loop filter is determinate of the maximum value for Y because the generated frequency should be filtered out. Based on empirical measurements and observations, it is apparent the best divider factors for the sine wave generator are: 4, 8, 16 or 32. Additionally, the amplitude of the sine wave signal can be varied, and it depends on the fractional input used as to how large an amplitude as a bit wide signal can be selected. The amplitude is a tradeoff between the shaping feature and output amplitude of the sine wave signal after passing through the loop filter. It is generally not possible to predict or know exactly the correct amplitude level for a given application. Accordingly, it is preferable to implement a number of different amplitude levels which are controlled and selected by a DSP to achieve the desired performance. Of course, the best choice is to use an amplitude level as small as possible to produce the needed noise shaping feature.

[0041] Turning now to FIG. 4, a delta-sigma fractional-N phase locked loop frequency synthesizer embodying the invention is illustrated therein and designated generally **300**. The delta-sigma fractional-N synthesizer **300** comprises a phase frequency detector (PFD) **302**, a loop filter **304**, a voltage-controlled oscillator (VCO) **306** and a dual-modulus frequency divider (DMD) shown in the dash line box **308** in a feedback loop between the VCO output **310** and an input **312** of the PFD **302**. A reference frequency F_{ref }is coupled to the input **314** of an R-divider designated generally **316**. An R-control signal on the R-input **318** causes the R-divider to be loaded with the desired count to scale the frequency F_{ref}. The R-divider functions and operates as described above in connection with FIG. 3. The output **320** of the R-divider **316** is coupled to one input **322** of the PFD **302**. The output frequency signal F_{out }on lead **310** produced by the VCO **306** is a fraction of the input frequency F_{ref }as input to the PFD **302** from the output **320** of the R-divider **316**. The VCO output **310** is coupled to the input **382** of the DMD **308**. The F_{comp }signal from the output **326** of the DMD **308** is coupled to a clock input **328** of a delta-sigma modulator (DSM) **330** and a clock signal input **332** of a sine wave generator **334** in addition to the input **312** of the PFD **302**.

[0042] The output **336** of the sine wave generator **334** is coupled to one input **338** of an adder **340** where the sine wave signal is combined with a fractional-N-bit control word F_{input }on the lead **342** coupled to the input **343** of the adder **340** to produce an “average zero” dither control signal at the adder output **344**. The output **344** is coupled to the input **346** of the DSM **330**. The DSM **330** produces a control signal at its output **348** which is coupled to one input **350** of an adder **352**. An A_{input }word on the lead **354** is coupled to an input **356** of the adder **352** and is added to the “dithered” control signal from the DSM output **348** to produce an “A” control signal at the adder output **358** and the adder output **362**. The output **358** from the adder **352** is coupled to the input **360** feeding the input **394** of the A-divider **380** of the DMD **308**. The “A” control signal from the adder output **358** controls the A-divider/counter **380** which determines how many times the prescaler **376** counts using P+1 clock signal. The output **362** of the adder **352** is coupled to the input **364** of adder **366**. An N-bit control word N_{input }on the lead **370** is coupled to the adder input **368** and is combined with the output **362** of adder **352** to produce an “N” control signal at the output **372** of the adder **366**. The “N” control signal is coupled to the input **374** which feeds the input **377** of the N-divider **378** of the DMD **308**. The “N” control signal at the input **374** carries information that controls the switching of the division ratio of the DMD **308**.

[0043] The DMD **308** comprises a prescaler **376**, an N-divider **378** and an A-divider **380**. The frequency F_{out }at the VCO output **310** is coupled to the input **382** of the prescaler **376**. The prescaler **376** is an integer divider controlled by the MOD signal presented at its input **398** from the output **396** of the A-divider **380** and divides by P or P+1 dependent on the value of the MOD signal. A scaled frequency F_{pre }of the VCO frequency F_{out }is generated at the output **384** of the prescaler **376** and is coupled to the input **386** of the N-divider **378** and to the input **388** of the A-divider **380** and functions as the clock signal to both dividers. The N-divider **378** generates a comparison frequency F_{comp }at the DMD output **326** in accordance with the control word at its input **377** fed from the input **374** and the prescaler frequency output F_{pre}. The output **390** of the N-divider **378** is coupled to the input **392** of the A-divider **380** to apply a division function to the prescaler frequency F_{pre }in accordance with the control word at the A-divider input **394**. The A-divider **380** produces a MOD control signal at its output **396** which is fed back to the input **398** of the prescaler **376** to cause the prescaler to change its division ratio. The output **390** of the N-divider is a load signal or a reset/set signal to the A-divider **380**. The output **390** is active during one F_{pre }clock period each time the N-divider **378** is full. This means that the count of the A-counter/divider **380** is always less than or equal to the count of the N-divider **378**. After the load signal the A-counter/divider **380** is set to zero and it counts until it reaches the value of the “A” control word at input **394**. Another approach other than loading the A-counter/divider **380** with a count of zero, is to set the A-counter/divider **380** to a maximum value minus the control word value, for example, (15−6=11 for 4 bit A-counter and a control word value 6). In this case, the A-counter output signal is used directly as the MOD signal. When the A-divider full signal is zero, the prescaler **376** counts using the P+1 clock signal and otherwise it counts using the P clock signal.

[0044] Turning now to FIG. 5, a delta-sigma fractional-N phase locked loop frequency synthesizer embodying the invention is illustrated therein and designated generally **400**. The delta-sigma fractional-N synthesizer **400** is particularly suited for use as an indirect delta-sigma fractional-N synthesizer or modulated fractional-N synthesizer, that is, the modulation is controlled by phase information. The delta-sigma fractional-N synthesizer **400** is similar to the embodiment illustrated in FIG. 4 and like reference numbers correspond to like components; however, in this case the F_{input }signal on lead **342** is the output **444** of the adder **470** and is the sum of the F_{ract }input signal coupled to the input **442** and the modulation data signal coupled to the input **454**. The delta-sigma fractional-N synthesizer **400** is also preferred in instances where the amplitude of the sine wave signal is of a magnitude that exceeds the amplitude that can be input to the delta-sigma modulator. The delta-sigma fractional-N synthesizer **400** comprises a phase frequency detector (PFD) **302**, a loop filter **304**, a voltage-controlled oscillator (VCO) **306** and a DMD shown in the dash line box **308** in a feedback loop between the VCO output **310** and an input **312** of the PFD **302**.

[0045] A reference frequency F_{ref }is coupled to the input **314** of an R-divider designated generally **316**. An R-control signal on the R-input **318** causes the R-divider to be loaded with the desired count to scale the frequency F_{ref}. The R-divider functions and operates as described above in connection with FIG. 3 and FIG. 4 above. The output **320** of the R-divider **316** is coupled to one input **322** of the PFD **302**. The output frequency signal F_{out }produced on lead **310** by the VCO **306** is a fraction of the input frequency F_{ref }appearing at the input to the PFD **302** from the output **320** of the R-divider **316**. The VCO output **310** is coupled to the input **382** of the DMD **308**. The F_{comp }signal from the output **326** of the DMD **308** is coupled to a clock input **328** of a delta-sigma modulator (DSM) **330**, a clock input **332** of a sine wave generator **334**, an input **414** of a first compensating delay means **402** and an input **420** of a second delay compensating means **404** in addition to the input **312** of the PFD **302**. In this embodiment it is critical to avoid delays in the output signal (clock cycles) of the DSM otherwise phase errors would be introduced which would change the control information to the A-counter/divider and N-counter/divider. Accordingly, any delays of the DSM clock cycles must be compensated before input to the dual-modulus frequency divider. Any delays in the clock cycles of the DSM depend on the specific implementation of the DSM. It should be noted that the delay compensating means are also clocked by the F_{comp }signal to maintain proper timing and may be implemented using registers or latches. A further possible implementation of the delay compensating means is a FIFO (first-in, first-out) structure using RAM cells and other clocked logic including the counter(s).

[0046] The output **336** of the sine wave generator **334** is coupled to one input **338** of an adder **340** where the sine wave signal is combined with a fractional N-bit control word F_{input }on the lead **342** coupled to the input **343** of the adder **340** to produce an “average zero” dither control signal at the adder output **344**. The output **344** is coupled to the input **346** of the DSM **330**. The DSM **330** produces a control signal (clock cycles) at its output **348** which is coupled to one input **410** of an adder **408**. The output **412** from the compensating delay means **402** is coupled to the input **426** of the adder **408** and the output **412** is added to the output **348** from the DSM **330**.

[0047] An A_{input }on the lead **354** is coupled to an input **356** of the adder **352** and is added to the sine wave+F_{input }“dithered” signal coupled to its input **350** from the output **345** of the adder **340**. The resultant sine wave+F_{input}+A_{input }at the output **362** of the adder **352** is coupled to the input **364** of the adder **366**. The output **353** of the adder **352** is coupled to the input **416** of the compensating delay means **402**. An N-bit word N_{input }on the lead **370** is coupled to the adder input **368** of adder **366** and is combined with the sine wave+F_{input}+A_{input }output **362** from the adder **352**. The resultant sine wave+F_{input}+A_{input}+N_{input }output **372** from the adder **366** is connected to the input **418** of the compensating delay means **404**. The output on lead **412** of the compensating delay means **402** is added in the adder **408** to the output **348** of the DSM to produce an “A” control signal at the output **428** which is input to the dual-modulus frequency divider **308** feeding the input **394** of the A-counter/divider. Another output **430** of the adder **408** is connected to an input **432** of an adder **406** where the signal at the output **430** is added to the signal at the output **422** from the compensating delay means **404** coupled to the input **424** of the adder **406**. The adder **406** produces an “N” control signal at the output **434** which is coupled to the input **374** to the dual-modulus frequency divider **308** feeding the N-counter/divider **378**. The adders **340**, **352** and **366** coupled to the F_{input}, A_{input }and N_{input}, respectively are used because the amplitude (weight or MSB) of the sine wave generator output signal may be same level as the N-input signal. An additional function of the adders **346**, **352** and **366** is to handle overflow and underflow situations in the calculations, for example, if the F_{input }is full 255 (8 bit), the A_{input }is full 15 (4-bit), the N_{input }is 23 (8 bit) and output of the sine wave generator is “0”, then everything is okay. However, if the output of the sine wave generator is “1”, then the output of the adder **340** is “0” and the output signal **345** goes “1”, causing the adder **352** to overflow. Now the output of the adder **352** is “0” and the output signal **362** is “1”, which is added together with the N input signal. Finally, the value of the output **372** of the adder **366** is 24.

[0048] As in FIG. 4, the DMD **308** in FIG. 5 comprises a prescaler **376**, an N-divider **378** and an A-divider **380**. The frequency F_{out }at the VCO output **310** is coupled to the input **382** of the prescaler **376**. The prescaler **376** is an integer divider controlled by the MOD signal presented at its input **398** from the output **396** of the A-divider **380** and divides by P or P+1 dependent on the value of the MOD signal. A scaled frequency F_{pre }of the VCO frequency F_{out }is generated at the output **384** of the prescaler **376** and is coupled to the input **386** of the N-divider **378** and to the input **388** of the A-divider **380** and functions as the clock signal to both dividers. The N-divider **378** generates a comparison frequency F_{comp }at the dual-modulus divider output **326** in accordance with the control word at its input **377** fed from the input **374** and the prescaler frequency output F_{pre}. The output **390** of the N-divider **378** is coupled to the input **392** of the A-divider **380** to apply a division function to the prescaler frequency F_{pre }in accordance with the control word at the A-divider input **394**. The A-divider **380** produces a MOD control signal at its output **396** which is fed back to the input **398** of the prescaler **376** to cause it to change its division ratio.

[0049] The sine wave signal used to “dither” the input to the delta-sigma modulator may be generated using any of a number of different techniques and methods now known or future developed, some of which methods are illustrated, for example, in FIGS. 6, 7 and **8**. In FIG. 6, the sine wave generator **334** comprises a counter **500** and a logic function means **502**. The compensating frequency F_{comp }is connected to the input **504** to the counter **500**. The counter **500** produces an N-bit word at its output **506**. The N-bit word is coupled to the input **508** of the logic function means **502** which, in response to the N-bit word input, produces an M-bit word sine wave signal at the output **510**.

[0050] In FIG. 7, the sine wave generator comprises a counter **500** and a read-only memory (ROM) **512**. The compensating frequency F_{comp }is connected to the input **504** to the counter **500**. The counter **500** produces an N-bit word at its output **506**. The N-bit word is coupled to the input **514** of the ROM **512**. An M-bit word is retrieved from an address location in the ROM corresponding to the N-bit word input and produces an M-bit word sine wave at the output **516**.

[0051] In FIG. 8, the sine wave generator comprises a counter **500** and a random access memory (RAM) **520**. The compensating frequency F_{comp }is connected to the input **504** to the counter **500**. The counter **500** produces an N-bit word at its output **506**. The N-bit word is coupled to the input **522** of the RAM **520** and is written into an address location when the RAM is enabled by a write signal on the write enable input **524**. An M-bit word is retrieved from the address location in the RAM corresponding to the N-bit word input and produces an M-bit word sine wave at the output **526**.

Referenced by

Citing Patent | Filing date | Publication date | Applicant | Title |
---|---|---|---|---|

US6870409 * | 8 May 2003 | 22 Mar 2005 | Samsung Electronics Co., Ltd. | Frequency synthesizer for reducing noise |

US7515931 * | 12 Dec 2005 | 7 Apr 2009 | Cisco Technology, Inc. | Frequency synthesizer and synthesis method for generating a multiband local oscillator signal |

US7911247 * | 26 Feb 2008 | 22 Mar 2011 | Qualcomm Incorporated | Delta-sigma modulator clock dithering in a fractional-N phase-locked loop |

US8406364 * | 20 Feb 2008 | 26 Mar 2013 | Fujitsu Semiconductor Limited | Fractional frequency divider PLL device and control method thereof |

US8742864 | 4 Nov 2010 | 3 Jun 2014 | Qualcomm Incorporated | Method and digital circuit for generating a waveform from stored digital values |

US20080198959 * | 20 Feb 2008 | 21 Aug 2008 | Fujitsu Limited | Fractional Frequency Divider PLL Device and Control Method Thereof |

WO2012061665A1 * | 4 Nov 2011 | 10 May 2012 | Qualcomm Incorporated | Method and digital circuit for generating a waveform from stored digital values |

Classifications

U.S. Classification | 331/100 |

International Classification | H03L7/197 |

Cooperative Classification | H03L7/1976, H03L7/1978 |

European Classification | H03L7/197D1, H03L7/197D1A |

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