EP0136818A1

EP0136818A1 - Dual mode feed horn or horn antenna for two or more frequency bands

Info

Publication number: EP0136818A1
Application number: EP84305895A
Authority: EP
Inventors: Charles M. Knop; Edward L. Ostertag
Original assignee: Andrew LLC
Current assignee: Commscope Technologies LLC
Priority date: 1983-09-06
Filing date: 1984-08-29
Publication date: 1985-04-10

Abstract

A microwave feed horn or horn antenna (11,12) for at least two frequency bands comprises a conical waveguide section (42) whose aperture at the large end has an inside diameter (D1) approximately equal to one wavelength in the lower frequency band so as to produce substantially equal power patterns in the E and H planes in the lower frequency band. The slope (ß) of the inside wall of said conical section (42) is selected to cancel the electric field at the aperture of the horn in the higher frequency band, thereby producing substantially equal power patterns in the E and H planes in the higher frequency band. A pair of straight waveguide sections (40,41) connects the opposite ends of the conical waveguide section (42).

Description

The present invention relates generally to microwave antennas and, more particularly, to feed horns or horn antennas that are capable of handling two or more frequency bands.
It is a primary object of the present invention to provide an improved feed horn or horn antenna that produces substantially equal E-plane and H-plane patterns in at least two different frequency bands, and yet is extremely simple and economical to manufacture. In this connection it is also an object of this invention to provide a feed horn or horn antenna suitable for simultaneous operation across two different frequency bands in both vertical and horizontal polarizations.
It is another important object of this invention to provide such an improved feed horn or horn antenna which is extremely small and, therefore, minimises the horn blockage of reflector-type antennas.
It is yet another object of this invention to provide an improved feed horn or horn antenna which achieves the foregoing objectives while maintaining a good VSWR (i.e. less than 1.1) and a low level of back radiation.
Other objects and advantages of the invention will become apparent from the following detailed description and the accompanying drawings.
In accordance with the present invention, the foregoing objectives are realised by a microwave feed horn or horn antenna for at least two frequency bands, the horn comprising a conical waveguide section whose aperture at the large end has an inside diameter approximately equal to one wavelength in the lower frequency band so as to produce substantially equal main beam patterns (from O to about 20 dB down) in the E and H planes in said lower frequency band, the slope of the inside wall of said conical section being selected to cancel the electric field at the inside wall of the horn at its large end in the higher frequency band, thereby producing substantially equal main beams in the E and H planes in said higher frequency band, and a pair of straight waveguide sections connected to opposite ends of said conical section.
In the drawings:

Fig. lA is a cross-sectional view of a dual-reflector antenna utilising the feed horn according to the invention;
Fig. 1B is a cross-sectional view of a parabolic antenna with a prime feed utilising the feed horn according to the invention;
Fig. 2 is an enlarged longitudinal section of the feed horn of the antenna of Fig. 1:
Fig. 3 is a plot of the radiation amplitude patterns, in both the E-plane and H-plane for the feed horn of Fig. 2, measured at a radius of 11" from the centre of the feed horn aperture and at a frequency of 3.95 GHz;
Fig. 4 is a plot of the radiation phase patterns in both the E-plane and the H-plane, for the feed horn of Fig. 2, measured at a radius of 11" from the centre of the feed horn aperture and at a frequency of 3.95 GHz;
Fig. 5 is a plot of the radiation amplitude patterns, in both the E-plane and the H-plane, for the feed horn of Fig. 2, measured at a radius of 11" from the centre of the feed horn aperture and at a frequency of 6.175 GHz; and
Fig. 6 is a plot of the radiation phase patterns, in both the E-plane and the H-plane, for the feed horn of Fig. 2, measured at a radius of 11" from the centre of the feed horn aperture and at a frequency of 6.175 GHz.

While the invention will be described in connection with certain preferred embodiments, it will be understood that it is not intended to limit the invention to those particular embodiments. On the contrary, it is intended to cover all alternatives, modifications and equivalents as may be included within the spirit and scope of the invention as defined by the appended claims.
Turning now to the drawings and referring first to Fig. lA, there is illustrated a dual-reflector Gregorian antenna comprising a paraboloidal main reflector dish 10, a primary feed horn 11 connected to and supported by a circular waveguide 12 extending along the axis of the dish 10, and a subreflector 13. The axis of the main dish 10 as shown in Fig. 1A is coincident with the longitudinal axis of the waveguide 12 and feed horn 11. (The term "feed" as used herein, although having an apparent implication of use in a transmitting mode, will be understood to encompass use in a receiving mode as well, as is conventional in the art).
In the transmitting mode, the feed horn 11 receives microwave signals via the circular waveguide 12 and launches those signals onto the subreflector 13; the subreflector 13 reflects the signals onto the main reflector dish 10, which in turn reflects the signals in a generally planar wave across the face of the paraboloid. In the receiving mode, the paraboloidal main reflector 10 is illuminated by an incoming planar wave and reflects this energy into a spherical wave to illuminate the subreflector 13; the subreflector 13 reflects this incoming energy into the feed horn 11 for transmission to the receiving equipment via the circular waveguide 12.
As is required in Gregorian dual-reflector antennas, the focal point F of the paraboloidal surface of the main reflector is located between the main reflector dish 10 and the subreflector 13. To achieve this configuration, the subreflector 13 presents a concave reflective surface to the face of the main reflector 10. To support the subreflector 13 in this desired position, the subreflector is mounted on the large end of a tripod 14 fastened to brackets 15 on the main reflector dish 10. The tripod is composed of three support legs which are relatively thin and introduce only a negligible amount of VSWR and pattern degradation into the antenna system. Normally the legs of the tripod are arranged to lie outside the horizontal plane. Alternatively, the subreflector can be supported by a dielectric cone with the small end of the cone mounted on the main reflector 10, or on the waveguide 12, and the subreflector mounted on the large end of the cone.
The subreflector 13 is positioned and dimensioned to intercept a large portion of the radiation launched from the feed horn 11 in the transmitting mode, and an equally large portion of the incoming radiation reflected by the main reflector 10 in the receiving mode, while at the same time minimizing blockage of the aperture of the main reflector 10. The subreflector preferably has a maximum diameter of about six wavelengths at the lowband frequency and nine wavelengths at the highband and is positioned sufficiently close to the feed horn to accomplish the desired interception of radiation from the horn.
In the illustrative embodiment, the subreflector 13 is fitted with an absorber-lined shield 30 which intercepts a substantial portion of the spillover from the feed horn 11 and also reduces diffraction of microwave radiation at the periphery of the subreflector 13. For the purpose of dissipating the spillover energy intercepted by the shield 30, the inner surface of this shield is lined with an absorber material 31. The shield 30 projects from the periphery of the subreflector 13 toward the main reflector and parallel to the axis of the feed horn. Since the Gregorian configuration of the antenna utilises a concave reflective surface on the subreflector (as contrasted with, for example, the convex reflective surface utilised in a Cassegrain configuration), the shield 30 can be added to the periphery of the subreflector 13 without interfering with the signal path between the subreflector 13 and the main reflector 10.
The axial length Ll of the shield 30 is limited by the surface of an imaginary cone whose apex is the common focal point F of the dual reflectors and whose base is the periphery of the main reflector (the cone surface is illustrated by the dotted line A-B, in Fig. lA). In three dimensions, this imaginary cone defines the surface within which the presence of the subreflector shield would interfere with the signal path between the main reflector 10 and the subreflector 13.
Diffraction normally occurs at an edge of a subreflector. However, with the addition of the subreflector shield 30, the only diffracting edge of the subreflector assembly, i.e., the edge of the shield 30, is located in a region where the spillover energy level is significantly less than at the periphery of the subreflector 13. As a consequence, the diffraction caused by the subreflector assembly with the shield 3₀ is much less than without the shield, producing lower side lobes in the region beyond about 10° off axis.
Referring to Fig. lA, the edge of the subreflector shield 30 is shown to be at an angle θ₂ with respect to the axis of the main dish shown in Fig. 1, while the edge of the subreflector 13 is at an angle 9₁ with respect to the axis of the main reflector. Since the radiation beam, as it leaves the feed horn 11, has its peak on the axis of the main reflector 10, the spillover energy level of the beam emanating from the feed horn 11 at angle 9₂ is significantly lower than it is at angle θ₁. Consequently, diffraction of that portion of the beam impinging on the periphery of the shield 30 (at angle 8₂) contributes substantially less to the side lobe patterns than would diffraction of the beam from the edge of the subreflector 13 (at angle θ₁), which corresponds to a higher energy level within the beam path. In other words, the addition of the shield 30 moves the diffracting edge of the subreflector assembly from the relatively high-energy angle 9₁ to the relatively low-energy angle θ₂.
To capture the spillover energy that is not intercepted by the subreflector shield 30, a shield 32 is provided on the main reflector 10. This shield 32, which has a relatively short axial length L2, is also lined with absorbing material 31. The lengths Ll and _L2 of the two shields 30 and 32 are such that their combined effect is to intercept and dissipate substantially all the spillover radiation from the feed horn 11. With these two shields 30 and 32, the antenna exhibits much improved RPE side lobes.
In order to minimize the size of the main reflector shield 32, the axial length Ll of the subreflector shield 30 is preferably maximized. The upper limit for the length Ll of the subreflector shield is the imaginary cone mentioned earlier, representing the outermost portion of the signal path between the two reflectors. In practice, the shield length Ll is made slightly shorter than its maximum permissible length to ensure that it does not interfere with the desired beam.
The shield 30 is positioned on the periphery of the subreflector 13. Any number of means for attaching the shield to the subreflector can be used, depending on the materials of construction used for the shield and subreflector. The shield is preferably constructed of a continuous flat metal or fiberglass projection in an annular shape whose inner and outer walls are substantially parallel to the axis of the subreflector. Conventional microwave absorbing material having a pyramidal, flat or convoluted surface, or even "hair" absorber, can be used on the inside surface of the shield.
The main reflector shield 32 is constructed in a manner similar to the subreflector shield 30. The shield 32 is also constructed of an annular metal or fiberglass projection whose inner and outer walls are substantially parallel to the axis of the main reflector. The inner wall can be lined with the same microwave absorbing material used in the subreflector shield 30. The bottom of the shield 30 is usually slightly shorter than the top so that the planar radome surface 17 is slightly tilted relative to the axis so as to improve the VSWR, i.e., reflections off the radome are not parallel to the axis.
_Fig. 1B shows the use of a feed horn lla, similar to the feed horn 11 to be described in more detail below, as a prime feed for a parabolic antenna having a main reflector dish lOa having a diameter D and a focal length F. As in the application Fig. lA, the preferred (though not necessary) choice of the subtended angle ψ_D from the centre of the horn aperture to the edge of the reflector lOa is about 52⁰, though this value can vary widely (e.g., from 45⁰ to 80⁰, approximately) depending upon the desired gain- versus-pattern trade-off. Since the F/D ratio of a paraboloidal dish is given by the equation F/D = 1/4tan (ψ_D/2), it follows that the choice of F/D should be about 0.51. Thus, the antenna of Fig. 1B will provide a paraboloidal dish antenna having substantially equal E and H plane patterns at two different bands of operation, simultaneously.
Referring next to Fig. 2, the feed horn 11 comprises two straight circular waveguide sections 40 and 41 interconnected by a conical circular waveguide section 42. Conical feed horns have been well known for a long time, and it has been known that the TM₁₁ mode can be excited in such horns by feeding a signal to the horn in the TE₁₁ mode. It has also been known that substantially equal E-plane and H-plane patterns can be produced by controlling the ratio of the mode powers in such horns.
However, the feed horn of this invention produces substantially equal E-plane and H-plane patterns not only in one, but in two different frequency bands. This is accomplished by selecting the diameter of the horn mouth to be approximately equal to one wavelength in the lower frequency band, and then selecting the slope of the conical wall to cancel the radial electric field at the aperture of the horn (of inner diameter Dl) in the upper frequency band. The one-wavelength diameter for the lower frequency band produces substantially equal patterns in the E and H planes for the lower-frequency signals, while the cancellation of the electric field of the higher-frequency signals at the inside wall of the horn aperture produces substantially equal patterns in the E and H planes for the higher-frequency signals. The horn is both small and inexpensive to fabricate, and yet it produces optimum main beam patterns in both the E and H planes in two different frequency bands simultaneously. The small size of the horn means that it minimizes horn blockage in prime-fed reflector-type antennas (of the type shown in Fig. 1B) and reduces horn shadow on the dish in the dual reflector antennas (of the type in Fig. 1A).
The illustrative feed horn 11 (Fig. 2) is a conventional smooth-wall TEll-mode horn at the low frequency (e.g., 3.95 GHz) with an inside diameter Dl in its larger cylindrical section 40 approximately equal to the wavelength at the center frequency (e.g., 3.95 GHz) of the lower frequency band. The second cylindrical section 41 of the feed hern has a smaller inside diameter D2, and the two cylindrical sections 40 and 41 are joined by the uniformly tapered conical section 42 to generate (at the junction of sections 40 and 42) and propagate the TM₁₁ mode in the upper frequency band (e.g., 6 GHz). More specifically, the conical section 42 generates (at the junction of sections 40 and 42) a TM₁₁ mode from the TEll mode propagating from left to right in the smaller cylindrical section 41 and in the section 42. At the end of the conical section 42 the freshly generated TM₁₁ mode leads the TEll mode by about 90° in phase. The slope of the conical section 42 determines the amplitude of the TM₁₁ mode signal, while the length L of the larger cylindrical section 40 determines the phase relationship between the two modes at the aperture of the feed horn.
Proper selection of the length L of the cylindrical section 40 of the feed horn 11 insures that the TM₁₁ and TEll modes are in phase at the feed horn aperture, in the upper frequency band. Also, good impedance matching is obtained, with the feed horn design of Fig. 2 having a VSWR of less than 1.1. The inside diameter of the waveguide 12 coupled to the small end of the feed horn is the same as that of the smaller cylindrical section 41. A pair of coupling flanges 43 and 44 on the waveguide and feed horn, respectively, fasten the two together by means of a plurality of screws 45.
To suppress back radiation at the low band (in the direction of the main dish) from the external surface of the horn 11, the open end of the horn is surrounded by a quarter-wave choke 46 comprising a short conductive cylinder 47, concentric with the horn 11, and a shorting ring 48. The inner surface of the cylinder 47 is spaced away from the outer surface of the horn 11 along a length of the horn about equal to a quarter wavelength (at the low band) from the end of the horn, and then the cylinder 47 is shorted to the horn 11 by the ring 48 to form a quarter-wave coaxial choke which suppresses current flow on the outer surface of the horn.
At the high frequency band (for which the free space wavelength is λ_H), back radiation is suppressed, and equal main beams are obtained in the E and H planes, by cancelling the electric field at the aperture boundary. To achieve this, the ratio of the mode powers W and WTE₁₁ must be:
where the guide wavelength of the TM₁₁ mode is
The guide wavelength of the TEll mode is
and
The relationship between the above mode power ratio, the diameter Dl at the large end of the conical section 42, and the half flare angle β (in degrees) of the conical section 42 is known to be given by the following equation:
Equating equations (1) and (5) yields:
To produce approximately equal E and H patterns in the low frequency band, the diameter Dl is made about equal to one wavelength, ÀL, at the midband frequency of the low band, i.e.:
Thus, equation (6) becomes:
Equation (₈) can then be solved for β :
This value of p results, at the high band, in cancellation of the electric field at the aperture boundary, which in turn results in approximately equal E and H patterns of the main beam radiated from the horn in the high frequency band.
To ensure that the TM₁₁ mode is generated at the junction between the cylindrical section 40 and the conical section 42, the diameter Dl must be such that the value of C, which is defined by equation (4) as
, is above the Eigen value of 3.83 for the TM₁₁ mode in the high frequency band. To ensure that only the TM₁₁ mode is generated, the diameter Dl must be such that the value of C is below the Eigen value of 5.33 for the TE₁₂ mode in the high frequency band. Thus, the value of C must be within the range of from about 3.83 to about 5.33. The symmetry of the cylindrical sections 40 and 41 and of the conical section 42 insures that the other higher order modes (TM_O1 and TE₂₁) which can also propagate for (C > 3.83) will not be excited. Since Dl is selected to be equal to one wavelength λ_L for the low frequency band, equation (4) gives:
and, therefore, the ratio λ_L/λ_H must be within the range of from about 3.83/π to about 5.33/π , which is 1.22 to 1.61.
Thus, the two-frequency bands must be selected to satisfy the above criteria. One suitable pair of frequency bands are 4GHz and 6GHz, because λ_L and Dl are 2.953 inches, λ_H is 1.969 inches, and λ_L/λ_H is 1.5. This value of the ratio λ_L/λ_H is, of course, within the prescribed range of 1.22 to 1.61.
In one working example of the invention, a feed horn, of the type shown in Fig. 2 had an inner diameter of 2.125 inches in its smaller cylindrical section 40 and 2.810 inches in its larger cylindrical section 41. The conical section 42 connecting the two cylindrical sections had a half-flare angle B (via equation (9) ) of 30° with respect to the axis of the feed horn. The axial length of the conical section was 0.593 inches.. The lengths of the two cylindrical sections 41 and 40 were 1.0 inches and 4.531 inches, respectively.
The working example described above produced the E-plane and H-plane power patterns shown in Figs. 3 and 5 at 3.95 GHz and 6.175 GHz, respectively. The power patterns in Figs. 3 and 5 represent amplitude in decibels along an arc length of a circle whose centre is coincident _'with the position of the outer of the aperture of the antenna and whose radius is 11 inches. This same.feed horn produced the E-plane and H-plane phase patterns shown in Figs. 4 and 6 at 3.95 _GHz and 6.175 GHz, respectively.
From Figs. 3 and 5 it can be seen that, at a given band, the patterns are virtually identical in the E and H planes, and the amplitude is sufficiently low at 55° off axis (which is the location of the edge of a typical subreflector) to ensure adequate total energy capture by the subreflector. As to the phase patterns shown in Figs. 4 and 6, it will be noted that these curves are relatively flat, in both the E and H planes, out to 55° off axis.
As can be seen from the foregoing detailed description, the feed horn or horn antenna of the invention is small, extremely simple and economical to manufacture, yet it provides simultaneous operation over two frequency bands in both vertical and horizontal polarizations with substantially equal amplitude E-plane and H-plane patterns in both frequency bands. Equal patterns are produced in the low frequency band by selecting the diameter of the horn mouth equal to approximately one wavelength in the low frequency band. In the high frequency band, the conical circular waveguide section is dimensioned such as to cause cancellation of the electric field at the wall of the conical section. As indicated by the foregoing equations (l)-(10), proper selection of the half flare angle / results in the cancellation of the electric field and, as a result, equal amplitude E-plane and H-plane patterns are generated in the high frequency band. The length L of the larger cylindrical section can be adjusted to provide a satisfactory in-phase condition between the two modes at the horn mouth in the upper frequency band.

Claims

1. A microwave feed horn or horn antenna for at least two frequency bands, said horn being characterised by the combination of

a conical waveguide section (42) whose aperture at the large end has an inside diameter (Dl) approximately equal to one wavelength at the midband frequency in the lower frequency band so as to produce substantially equal main beams in the E and H planes in said lower frequency band,

the slope (β) of the inside wall of said conical section (41) being selected to cancel the electric field at the inside wall of the horn aperture in the higher frequency band, thereby producing substantially equal main beams in the E and H planes in said higher frequency band, and

a pair of straight waveguide sections (40,41) connected to opposite ends of said conical section (42).

2. A microwave feed horn or horn antenna as claimed in claim 1, characterised in that the small end of said conical waveguide section (42) has a diameter (D2) small enough to prevent propagation of the TM11 mode of microwave signals in both of said-frequency bands.

3. A microwave feed horn or horn antenna as claimed in either preceding claim, characterised in that the length (L) of the straight waveguide section (40) connected to the large end of said conical waveguide section (42) is selected to produce in-phase TEll and TM₁₁ modes of microwave signals in said upper frequency band at the open end of said straight waveguide section (40).

4. A microwave feed horn or horn antenna as claimed in any preceding claim, characterised in that the diameter (Dl) at the large end of said conical waveguide section (42) is large enough to permit the propagation of the TM₁₁ mode of microwave signals in said upper frequency band, and small enough to prohibit the propagation of the TE₁₂ mode of such signals.

5. A microwave feed horn or horn antenna as claimed in any preceding claim, characterised in that the inside diameter (Dl) at the large end of said conical waveguide section (42) yields an Eigen value within the range of from about 3.83 to about 5.33 at said higher frequency band, so that the TM₁₁ mode, and only the TM₁₁ mode, of signals in said higher frequency band is generated at the large end of said conical waveguide section (42).

6. A microwave feed horn or horn antenna as claimed in any preceding claim, characterised in that the ratio of the wavelength at the midband frequency in said lower frequency band to the wavelength at the midband frequency in said higher frequency band is within the range of from about 1.22 to about 1.61.

7. A microwave feed horn or horn antenna as claimed in any preceding claim, characterised in that said conical waveguide section (42) has a uniform slope β (in degrees) defined by the equation

where ÀH is the wavelength at the midband frequency of said higher frequency band, and AL is the wavelength at the midband frequency of said lower frequency band.

8. A microwave feed horn or horn antenna for two or more frequency bands, said horn being characterised by

a first waveguide section (40) forming the open end of the horn and having an inside dimension (Dl) which is approximately equal to the wavelength of a microwave signal at the midband frequency in the lower of said frequency bands so as to produce substantially equal E-plane and H-plane patterns for signals in said lower frequency band,

a second waveguide section (41) forming another portion of the horn and having an inside dimension (D2) smaller than that of said first waveguide section (40), and

a tapered waveguide section (42) joining said first and second sections (40,41) and having a slope (fi) which produces substantially equal E-plane and H-plane patterns for signals in said higher frequency band.

9. A microwave antenna characterised by the combination of:

a paraboloidal main reflector (10) having a focal point F;

a subreflector (13) forming a surface of revolution about the axis of said main reflector (10) and having a focal point between said main reflector (10) and said subreflector (13) and substantially coincident with the focal point (F) of said main reflector (10),

a feed horn (11,12) extending along the common axis of said main reflector (10) and said subreflector (13) for transmitting microwave radiation to, and receiving microwave radiation from, said subreflector (13) along a feed horn beam path, said feed horn (11,12) comprising

a conical waveguide section (42) whose aperture at the large end has an inside diameter (D1) approximately equal to one wavelength at the midband frequency in the lower frequency band so as to produce substantially equal patterns in the E and H planes in said lower frequency band,

the slope (β) of the inside walls of said conical horn (42) being selected to cancel the electric field at the horn aperture in the higher frequency band, thereby producing substantially equal patterns in the E and H planes in said higher frequency band.

10. A microwave antenna characterised by the combination of:

a paraboloidal reflector (10) having a focal point F,

a feed horn (11,12) extending along the axis of said reflector (10) for transmitting microwave radiation to, and receiving microwave radiation from, said reflector, the centre of the aperture of said feed horn (11,12) being located at the focal point (F) of said reflector (10), said feed horn (11,12) comprising

a conical waveguide section (42) whose aperture at the large end has an inside diameter (Dl) approximately equal to one wavelength at the midband frequency in the lower frequency band so as to produce substantially equal patterns in the E and H planes in said lower frequency band,