CA2251946A1 - Digital radio frequency interference canceller - Google Patents

Digital radio frequency interference canceller Download PDF

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Publication number
CA2251946A1
CA2251946A1 CA002251946A CA2251946A CA2251946A1 CA 2251946 A1 CA2251946 A1 CA 2251946A1 CA 002251946 A CA002251946 A CA 002251946A CA 2251946 A CA2251946 A CA 2251946A CA 2251946 A1 CA2251946 A1 CA 2251946A1
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Canada
Prior art keywords
frequency
interference
frequency domain
samples
radio
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Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Abandoned
Application number
CA002251946A
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French (fr)
Inventor
Brian R. Wiese
John A. C. Bingham
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Texas Instruments Inc
Original Assignee
Brian R. Wiese
Amati Communications Corporation
John A. C. Bingham
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Application filed by Brian R. Wiese, Amati Communications Corporation, John A. C. Bingham filed Critical Brian R. Wiese
Publication of CA2251946A1 publication Critical patent/CA2251946A1/en
Abandoned legal-status Critical Current

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Classifications

    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/26Systems using multi-frequency codes
    • H04L27/2601Multicarrier modulation systems
    • H04L27/2647Arrangements specific to the receiver only
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/10Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/12Neutralising, balancing, or compensation arrangements
    • H04B1/123Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/10Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/12Neutralising, balancing, or compensation arrangements
    • H04B1/123Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means
    • H04B1/126Neutralising, balancing, or compensation arrangements using adaptive balancing or compensation means having multiple inputs, e.g. auxiliary antenna for receiving interfering signal
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L27/00Modulated-carrier systems
    • H04L27/26Systems using multi-frequency codes
    • H04L27/2601Multicarrier modulation systems
    • H04L27/2602Signal structure
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00Arrangements affording multiple use of the transmission path
    • H04L5/003Arrangements for allocating sub-channels of the transmission path
    • H04L5/0044Arrangements for allocating sub-channels of the transmission path allocation of payload
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00Arrangements affording multiple use of the transmission path
    • H04L5/003Arrangements for allocating sub-channels of the transmission path
    • H04L5/0058Allocation criteria
    • H04L5/0062Avoidance of ingress interference, e.g. ham radio channels
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/10Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/1027Means associated with receiver for limiting or suppressing noise or interference assessing signal quality or detecting noise/interference for the received signal
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/10Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/1027Means associated with receiver for limiting or suppressing noise or interference assessing signal quality or detecting noise/interference for the received signal
    • H04B1/1036Means associated with receiver for limiting or suppressing noise or interference assessing signal quality or detecting noise/interference for the received signal with automatic suppression of narrow band noise or interference, e.g. by using tuneable notch filters
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04BTRANSMISSION
    • H04B1/00Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/06Receivers
    • H04B1/10Means associated with receiver for limiting or suppressing noise or interference
    • H04B1/1027Means associated with receiver for limiting or suppressing noise or interference assessing signal quality or detecting noise/interference for the received signal
    • H04B2001/1063Means associated with receiver for limiting or suppressing noise or interference assessing signal quality or detecting noise/interference for the received signal using a notch filter
    • HELECTRICITY
    • H04ELECTRIC COMMUNICATION TECHNIQUE
    • H04LTRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
    • H04L5/00Arrangements affording multiple use of the transmission path
    • H04L5/0001Arrangements for dividing the transmission path
    • H04L5/0003Two-dimensional division
    • H04L5/0005Time-frequency
    • H04L5/0007Time-frequency the frequencies being orthogonal, e.g. OFDM(A), DMT

Abstract

Disclosed are radio frequency (RF) interference cancellation techniques that effectively estimate RF interference to the data signals being received using a frequency domain model, and then remove the estimated RF interference from the received data signals. Improved techniques for digitally filtering multicarrier modulation samples to reduce sidelobe interference due to the RF
interference are also disclosed.

Description

CA 022~1946 1998-10-19 W O 97/40608 rCTrUS97/06384 DIGITAL RADIO FREQUENCY INTERFERENCE CANCF,T,~ R

CROSS-REFERENCE TO RELATED APPLICATIONS
This application incorporates by referenee the following International Patent Applications:
( I ~ PCT/US97/ ~ filed April 17, 1997, entitled "Radio Frequency Noise Canceller"
(claiming priority of U.S. Application No. 60/016,251, filed April 19, 1996), and (2) PCT/US97/ ~ filed April 17, 1997, entitled "Mitigating Radio Frequency Interference in Discrete Multicarrier Transmissions Systems", (claiming priority of U.S. Application No.
60/016,252.

BACKGROUND OF THE INVENTION
1. Field of the Invention The present invention rcla~es to radio frequency (RF) interference cancellation, and more particularly, to RF interference cancellation in multicarrier transmission systems.
s 2. Des~ noftheRelatedArt Bi-directional digital dat;l ~r;msmission systems are presently being developed for high-speed data communication. Onc standard for high-speed data communications over twisted-pair phone lines that has develo~d is known as Asymmetric Digital Subscriber Lines (ADSL).
Another standard for high-speed d;lt;l communications over twisted-pair phone lines that is 20 presently proposed is known ;~i Vcry Hi~h Speed Digital Subseriber Lines (VDSL).
The Alliance For Tclccommunic;~tions Information Solutions (ATIS), whieh is a group accredited by the ANSI (Amcrlc;~n N;ltlonal Standard Institute) Standard Group, has finAli~ed a discrete multi tone based appro;lch ~or thc trAnsmiccion of digital data over ADSL. The standard is intended primarily for trarlsmi~lln~ vidco data and fast Internet access over ordinary telephone 2s lines, although it may be uscd in ;1 ~ ~ricty of other applications as well. The North American Standard is referred to as the ANSI T 1.413 ADSL Standard (hereinafter ADSL standard).
TrAn.cmi.c.cion rates under the ADSL standard are intended to facilitate the tran.cmiccion of inforrnation at rates of up to 8 million bits per second (Mbits/s) over twisted-pair phone lines.
The standardized system defines the use of a discrete multi tone (DMT) system that uses 256 30 "tones" or "sub-channels" that are each 4.3125 kHz wide in the forward (downstream) direction.
In the context of a phone system, the downstream direction is defined as tr~ncmicsions from the I

CA 022~1946 1998-10-19 ~0 97/40608 PCTAUS97/06384 central office (typically owned by the telephone company) to a remote location that may be an end-user (i.e., a residence or business user). In other systems, the number of tones used may be widely varied. However when modulation is performed efficiently using an inverse fast Fourier transform (I~l~ l ), typical values for the number of available sub-channels (tones) are integer 5 powers of two, as for example, 128, 256, 512, 1024 or 2048 sub-channels.
The ADSL standard also defines the use of a reverse signal at a data rate in the range of 16 to 800 Kbit/s. The reverse signal collGspollds to transmission in an up~llea,ll direction, as for example, from the remote location to the central office. Thus, the term ADSL comes from the fact that the data tr~ncmiccion rate is substantially higher in the downstream direction than in the lo upstream direction. This is particularly useful in systems that are intended to transmit video prog~d~ lir'g or video conferencing information to a remote location over telephone lines.
Because both downstream and upstream signals travel on the same pair of wires (that is, they are duplexed) they must be separated from each other in some way. The method of duplexing used in the ADSL standard is Frequency Division Duplexing (FDD) or echo canceling.
15 In frequency division duplexed systems, the upstream and downstream signals occupy different frequency bands and are separated at the transmitters and receivers by filters. In echo cancel systems, the upstream and downstream.signals occupy the same frequency bands and are separated by signal processing.
ANSI is producing another standard for subscriber line based tr~ncmiscion system, which 20 is referred to as the VDSL standard. The VDSL standard is intended to facilitate transmission rates of at least 12.98 Mbit/s and up to 51.92 Mbit/s or greater in the downstream direction. To achieve these rates, the tr~ncmission distance over twisted-pair phone lines must generally be shorter than the lengths permitted using ADSL. Simultaneously, the Digital, Audio and Video Council (DAVIC) is working on a similar system, which is referred to as Fiber To The Curb 25 (FTTC). The tr~n.cmi.csion medium from the "curb" to the customer premise is standard unshielded twisted-pair (UTP) telephone lines.
A number of modulation schemes have been proposed for use in the VDSL and FTTC
standards (hereinafter VDSL/FTTC). Most of the proposed VDSL/FTTC modulation schemes utilize frequency division duplexing of the upstream and downstream signals. Another promising 30 proposed VDSL/FTTC modulation scheme uses periodic synchronized upstream and downstream communication periods that do not overlap with one another. That is, the upstream and downstream communication periods for all of the wires that share a binder are synchronized.
With this arrangement, all the very high speed tran.cmiccions within the same binder are synchronized and time division duplexed such that downstream communications are not CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 tr~n~mitted at times that overlap with the tr~n.cmi~ion of upstream eommunieations. This is also referred to as a (i.e. "ping pong") based data tr~n.cmicsion seheme. Quiet periods, during which no data is transmitted in either direetion, separate the upstream and downstream eo~ unieation periods. For example, with a 20-symbol ~ù~c.rl~lllle, two of the DMT symbols in the 5 supelrl~llle are silent (i.e., quite period) for the purpose of faeilitating the reversal of tr~n~mi~cion direetion on the phone line. In sueh a ease, reversals in transmic.cion direetion will occur at a rate of about 4000 per second. For example, quiet periods of about 10-25 ,~LS have been proposed.
The synchronized approaeh ean be used a wide variety of modulation schemes, ineluding multi-earrier tran.cmiccion sehemes such as Discrete Multi-Tone modulation (DMT) or Discrete o Wavelet Multi-Tone modulation (DWMT), as well as single earrier tr~n~mi.c.~ion sehemes sueh as Quadrature Amplitude Modulation (QAM), Carrierless Amplitude and Phase modulation (CAP), Quadrature Phase Shift Keying (QPSK), or vestigial sideband modulation. When the synehronized time division duplexed approach is used with DMT it is referred to as synehronized DMT (SDMT).
A eommon feature of the above-mentioned tr~n.cmicsion systems is that twisted-pair phone lines are used as at least a part of the transmission medium that eonnects a central office (e.g., telephone company) to users (e.g., residence). It is difficult to avoid twisted-pair wiring from all parts of the interconnecting tr~n.cmi~.cion medium. Even though fiber optics may be available from a central offiee to the curb near a user's residenee, twisted-pair phone lines are used to bring in the signals from the curb into the user's home or business.
Although the twisting of the twisted-pair phone lines provides some proteetion against external radio interference, some radio interference is still present. As the frequency of transmission inereases, the radio interference that is not mitigated by the twisting beeomes substantial. As a result, the data signals being transmitted over the twisted-pair phone lines at high speeds ean be signifieantly degraded by the radio interference. As the speed of the data transmission inereases, the problem worsens. For example, in the ease of VDSL signals being transmitted over the twisted-pair phone lines, the radio interferenee ean eause signifieant degradation of the VDSL signals. This problematic radio interference is also referred to as radio frequency noise.
The undesired radio hlLe.r~r~i,ee ean eome from a variety of sourees. One partieular souree of radio interference is amateur (or ham) radio operators. Amateur radios broadcast over a wide range of frequeneies with signifieant amount of power. The amateur radio operators also tend to change their broadcast frequency quite often, for example, about every two minutes.
Another souree of radio interferenee is AM radio transmissions by radio stations whieh broadeast CA 022~1946 1998-10-19 Wo 97/40608 PCT/US97/06384 over a wide range of frequencies. With high speed data tr~n.cmi.c.cion, the radio interference (noise) produced by various sources can significantly degrade the desired data signals being transmitted over twisted-pair phone lines.
Consequently, the problem with using twisted-pair phone lines with high frequency data 5 tr:ln.~mi~ion rates, such as available with ADSL and VDSL, is that radio interference becomes a substantial impediment to a receiver being able to be pl~e.ly receive transmitted data signals.
Thus, there is a need to provide techniques to elimin~t~ or compensate for radio interference.

SUMMARY OF THE INVENTION
o Broadly speaking, the invention pertains to radio frequency (RF) intelre~ ce cancellation techniques that effectively estimate RF interference to transmitted data signals being received using a frequency domain model for the RP interference, and then remove the estim~t~d RF
in~lrelcn~ce from the received data signals. The invention also pertains to improved techniques for digitally filtering multicarrier modulation samples to reduce sidelobe interference due to the s RF interference.
The invention can be implemented in numerous ways, including as an app~Lus, system, method, or computer readable media. Several embodiments of the invention are discussed below.
As a method for mitigating radio frequency (RE~) interference in a multicarrier modulation system, one embodiment of the invention includes the operations of: obtaining frequency domain data associated with a frequency band; identifying a restricted frequency sub-band within the frequency band; estim~ting a frequency of the RF interference within the restricted frequency sub-band; estim~ting the RF h~lelrelcnce in accordance with a frequency domain model for the RF
interference and the estim~t~d frequency of the RF interference; and thereafter removing the estim:~t~l RF interference from the frequency domain data.
As a method for mitigating radio frequency interference in a multicarrier modulation system, another embodiment of the invention includes the operations of: identifying AM radio clrelcnce to the multicarrier modulation system, estimating a frequency of the AM radio inL~lrercnce, and disabling certain frequency tones of the multicarrier modulation system adjacent to the estimated frequency of the AM radio interference from carrying data during the data transmission, these operations occur prior to data tr~n.~mi~sion. Thereafter, during or following data reception, the invention also includes the operations of estim~ting the AM radio inte~rc.ellce in accordance with a frequency domain model for the AM radio interference and the estim~t~d CA 022~1946 1998-10-19 wo 97/40608 PCT/US97/06384 frequency of the AM radio interference, and removing the estimated AM radio il.lclre~ ce from the frequency domain data on those of the frequency tones of the mllltic~rrier modulation system that carry data.
As a method for digitally filtering multicarrier modulation samples to reduce sidelobe 5 intelrtlcnce from a radio frequency (RF) interferer, the multicarrier modulation samples occur at predetermined frequency tones and form a multicarrier modulation symbol, an embodiment of the invention includes the operations of: receiving x samples of a multicarrier modulation symbol and y samples of a cyclic prefix associated with the multicarrier modulation symbol, the y samples of the cyclic prefix preceding the x samples of the multicarrier modulation symbol;
0 discarding an initial portion of the y samples of the cyclic prefix associated with the multicarrier modulation symbol; storing a remaining portion of the y samples of the cyclic prefix associated with the multicarrier modulation symbol; retaining a first portion of the x samples of the multicarrier modulation symbol without modification; and modifying a second portion of the x samples of the multicarrier modulation symbol in accordance with the stored samples of the 5 remaining portion of the y samples of the cyclic prefix and predetermined multiplication coefficients.
As a receiver for a multicarrier modulation system, an embodiment of the invention includes: an analog-to-digital (A/D) converter, a multicarrier demodulator operatively connected to the A/D converter, and a digital RF interference canceller operatively coupled to the multicarrier 20 demodulator. The AtD converter receives analog signals that have been transmitted to the receiver over a tr~n~mi~.~ion media and converts the analog signals to digital time domain signals. The multicarrier demodulator receives the digital time domain signals and converts the digital time domain signals into digital frequency domain data. The digital RF interference canceller mitigates the effect of RF interference on the digital frequency domain data by modeling the RF interference 25 in accordance with a frequency domain model. Preferably, the digital time domain signals include a plurality of multicarrier modulation symbols carrying data, with each of the symbols also including a guard band, and the receiver further includes a cyclic prefix removal and windowing processor operatively connected between the A/D converter and the multicarrier demodulator.
The cyclic prefix removal and windowing processor performs a time domain windowing 30 operation on the symbols.
Other aspects and advantages of the invention will become apparent from the following detailed description, taken in co~junction with the acc-~---pal-ying drawings, illustrating by way of example the principles of the invention.

CA 022~1946 1998-10-19 wo 97/40608 PCT/US97/06384 BRIEF DESCRIPTION OF THE DRAWINGS

The present invention will be readily understood by the following detailed description in conjunction with the accompanying drawings, wherein like reference numerals designate lilce structural elements, and in which:
FIG. l is a block diagram of a representative teleco~ lnirations system suitable for using the invention;
FIG. 2 is a graph illustrating a proposed trAn~mi~ion power spectral density forVDSL/FTTC upstream communications;
o FIG. 3 is a graph illustrating the magnitude of the maximum in-tone received power at a remote unit as a function of transmission frequency in a typical VDSL application over twisted-pair transmission lines;
FIG. 4 is a graph illustrating the magnitude of the maximum in-tone received power at a remote unit as illustrated in FIG. 3 further taking into consideration the effects turning off the 15 tones in restricted frequency bands;
FIG. S is a diagram illustrating frequency tones of a multicarrier modulation system having radio interference in a restricted frequency band;
FIG. 6 is a diagram illustrating the amount of radio interference ind~1ce~1 by a radio interferer on various frequency tones of a multicarrier modulation system;
FIG. 7 is a block diagram of a receiver for a multicarrier modulation system according to an embodiment of the invention;
FIGs. 8A-8C are diagrams illustrating various time domain models that modulate asinusoid;
FIG. 9 is a diagram of basic radio frequency (RF) cancellation processing according to a 25 basic embodiment of the invention;
FIGs. l0A and l0B are flow diagrams of digital RF cancellation processing according to an embodiment of the invention;
FIG. l l is a flow diagram of AM radio frequency (RF) cancellation processing according to an embodiment of the invention;
FIG. 12 is a flow diagram of prefix removal and windowing processing according to an embodiment of the invention; and FIG. l 3 is a diagram illustrating a 512 sample DMT symbol l 300 with a 40 sample prefix l 302, and a non-rectangular, extended window.
-CA 022~1946 1998-1o-19 ~V097/40608 PCT~US97/06384 DETATT,F,n DESCRIPTION OF THE INVENTION
In a multic~rrier modulation system using wideband mlllti~rrier modulation, radio frequency (RF) h~t~lrelcnce can often prevent proper reception of data llallsll~ilted by the multicarrier modulation system. The invention provides improved techniques for cancelling RF
- 5 hlte,re-c~nce, particularly from narrowband interferers, from the data transmitted by the multicarrier modulation system. More particularly, the invention pertains to radio frequency (RF) h~ ce cancellation techniques that effectively estimate RF interference to transmitted data signals being received using a frequency domain model, and then remove the estimated RF
interference from the received data signals. The invention also pertains to improved techniques 0 for digitally filtering multicarrier modulation samples to reduce sidelobe interference due to the RF interference.
Embodiments of the invention are discussed below with l~re.~llce to FIGs. 1-12.
However, those skilled in the arl will readily appreciate that the detailed description given herein with respect to these figures i.s for cxplanatory purposes as the invention extends beyond these 5 limited embodiments.

FIG. I is a block di~c~ram of a representative telecommunications system 2 suitable for using the invention. The telecommunications system 2 represents portions of a typical wired telecommunications system lhc~l is suitable for the VDSL and FTTC (hereinafter VDSL/FTTC) applications. The system ' includes a central office 10 that services a plurality of distribution 20 posts which may take the form of optical network units (ONUs) I l. Each distribution post communicates with the cen(rill of rlce 10 over one or more high speed, multiplexed transmission lines 12 (e.g., a fiber optic linc). Thc ONU 11 typically serves a multiplicity of discrete subscriber lines 15. Each subscribcr linc 15 typically services a single end user that is located within 1.5 kilometers of the Ol~ . l h~ end user would have a remote unit 18 suitable for 25 co,,,,,,lll-ic~ing with thc O~l~' I I ;It very high data rates. The remote unit 18 includes a modem but may take the form of ~ ncl! Or different devices, as for example, a telephone, a television, a monitor, a conlpu~r, a confcrcnclnc~ uni(, etc. Of course, it is possible that the end user may have a plurality of phones or other r mole unils 18 wired to a single line. The subscriber lines 15 serviced by a single ONU lypically leave the ONU 11 in a shielded binder 21. The shielding in 30 the binder generally serves as a good insulator against the emission (egress) and reception (ingress) of RF noise. However, the last segment of this subscriber line, commonly referred to as a "drop" 23 branches off from the binder and is coupled directly or indirectly to the end user's remote unit 18. This "drop" 23 portion of the subscriber line 15 between the remote unit 18 and the binder 21 is typically unshielded. In most applications the length of the "drop" is not more 35 than about 30 meters. However, the unshielded wire of the "drop" 23 effectively acts as an CA 022~1946 1998-10-19 ~nt~.nn:~ that both emits and receives RF signals. Additionally, there is some concern that the connection 25 between the ONU 11 and the twisted-pair subscriber lines 15 may also serve both as an RF energy emission source and as the receptor of RF energy.

The amount of energy that a particular co""l~ln~ication system may transmit is regulated 5 by both governmental and practical considerations. As intli~t.od above, in discrete multi-tone systems suitable for use in the VDSL/FTTC applications, frequency bands on the order 12 MHz are being contemplated. Within that 12 MHz frequency range, there are several narrow bands that are allocated to ~m~te~lr radio users. Thus, one proposed transmission power spectral density for VDSL/FTIC upstream communications is illustrated in FIG. 2. In this embodiment the transmit 0 power mask permits a maximum of -60 dBmJHz throughout the majority of the frequency band.
However, in selected frequency bands where amateur radio RF hlle,r~c"ceis expected (i.e., 1.8 to 2.0 MHz, 3.5 to 4.0 MHz, 7.0 to 7.3 MHz, and 10.1 to 10.15 MHz) tr~n.~mi~ions are limited to significantly lower levels. The permissible output power level in these restricted frequency bands varies somewhat between proposals. However, most parties to the VDSL/FTTC
5 standardization process have proposed maximum power densities in the range of approximately -70 dBm/Hz to -85 dBm/Hz. Regardless of the actual transmission power that is eventually agreed upon, it is clear that a conscious effort needs to be made to minimi7e emissions in the prohibited ranges.

A number of multi-carrier modulation schemes have been proposed for use in the VDSL
20 and FTTC standards (hereinafter VDSL/FTTC). One proposed multi-carrier solution utilizes discrete multi-tone signals in a system that is similar in nature to the ADSL standard. Other proposed modulation schemes include carrierless amplitude and phase moduiated (CAP) signals and discrete wavelet multi-tone rnodulation (DWMT). In order to achieve the data rates required by VDSL/FTTC, the tr~n.cmi~.~ion bandwidth must be significantly broader than the bandwidth 25 contempl~t~cl by the ADSL. By way of example, the discrete multi-tone system adopted for ADSL applications utilizes a transmission bandwidth on the order of 1.1 MHz, while bandwidths on the order of 12 MHz are being collte"lplated for VDSL/FTTC applications. In one proposed DMT system for VDSL/FTTC applications, the use of 256 "tones" or "sub-channels" that are each 43.125 kHz wide is contemplated.

As will be appreciated by those skilled in the art, high frequency multi-carrier signals transmitted over twisted-pair tr:~n~mi.c~ion lines experience significant ~ttenll~tion when they are transmitted a relatively long distance over the twisted-pair lines. FIG. 3is a graph illustrating the magnitude of the maximum in-tone received power at a remote unit (e.g., receiver) as a function CA 022~1946 1998-lo-19 W O 97/40608 PCTrUS97/06384 of tr~n.cmi~.cion frequency in a typical VDSL application over twisted-pair transmission lines. By way of example, referring to FIG. 3, when the transmit power is on the order of -60 dBm/Hz throughout the tr~n.cmi.~.cion bandwidth of a DMT based VDSL modulation scheme, the receive power at a typical remote user may be on the order of -70 dBrnlHz at the lower end of the 5 frequency spectrum, but may drop to as low as -125 dBmlHz at the higher end of the frequency spectrum. Thus, in situations where the "drop" 23 is located relatively far from the source, the - downstream signals may be attenuated enough by the time they reach the ~'drop" 23 that they are already below the permissible power spectral density. FIG. 4 is a graph illustrating the magnitude of the maximum in-tone received power at a remote unit as illustrated in FIG. 3 further taking into 0 consideration the effects of turning off the tones in the restricted frequency bands.

In any event, in multi-carrier tr;ln~mi~.~ions schemes such as DMT, there will naturally be a number of subcarriers (tones) that fall within the restricted frequency bands. Accordingly, a first step in reducing transmissions in the restricted frequency bands is to turn off those particular subcarriers. This has the advantage of both reducing the emissions in the prohibited frequency s range as well as reducing the adverse impacts associated with ingress (receipt) of the radio signals. However, as will be appreciated by those skilled in the art, it is difficult to contain the amount of power emitted for a particular tone tightly around a desired frequency center (fc ) Emissions associated with a particular tone typically include a relatively high power emission centered about the frequency center (fc ) and a number of side lobes of decreasing intensity 20 extending on either side thereon.

The magnitude and phase of the sidelobe power can make it difficult to limit the power spectral density in a narrow range within the DMT tr~n.cmi~sion band by simply turning off the tones within the restricted frequency band. By way of example, consider a system which uses tones that are 43.125 kHz wide. If an attempt is made to form a 200 kHz wide notch in the 1.8 to 2s 2.0 MHz range by simply turning off the tones within the 200 kHz wide prohibited range, the emission power at the center of the prohibited range would only be reduced from -60 dBm/Hz to on the order of -73 dBmlHz. Obviously, this might result in emissions above the desired range of -70 or -85 dBmlHz even in the center of the prohibited frequency range. Of course, the emission power at frequencies closer to the boundaries of the prohibited frequency range would 30 be significantly higher. Thus, if an attempt is made to reduce the emissions simply by turning off a range of tones in the multi-carrier transmissions system, the number of tones that need to be turned off would be significantly higher than the number of tones associated with a prohibited frequency range. Although the discrete multi-tone system is very flexible in its ability to pick and choose the subcarrier frequencies, the requirement of turning off such large frequency bands to CA 022~1946 1998-10-19 ~Vo 97/40608 PCTIUSg7/06384 avoid amateur radio int~r~lGnce is undesirable and may reduce system performance. Improved techniques for re~lncing RF emissions in restricted frequency bands are described in lnternational Patent App}ication No. PCT/US97/ filed April 17, 1997, entitled "Mitigating Radio Frequency Interference in Discrete Multicarrier Transmissions Systems" which has been incorporated by reference.

The invention primarily concerns the ingress of RF inte~r~,lG~Ice (RP energy) into twisted-pair tr~ncmi~.cion lines (e.g., "drop" 23). The RF interference may be from a variety of different RF interference sources, including an amateur radio opel~tOr and AM radio stations. According to the invention, the RF inl~l~el~nce is able to be located, estimated and cancelled from data o signals being received.

FIG. 5 is a diagram 500 illustrating frequency tones of a multicarrier modulation system having radio interference in a restricted frequency band. As an example, the multicarrier modulation system may be a Discrete Multi-Tone (DMT) modulation system. The diagram 500 is a frequency domain illustration of a plurality of tones 502 on predetermined frequencies of the s multicarrier modulation system. Data information is transmitted on the tones 502. However, the frequencies over which the tones 502 are able to be transmitted often include one or more restricted frequency bands in which data should not be transmitted 504. However, radio interference is often produced in the restricted frequency band 504 because of radio tr~n.~mi~.cions by others. As an example, in the restricted frequency band 504 illustrated in FIG. 5, a radio 20 interferer 506 transmits within the restricted frequency band 504. The radio interferer 506, for example, could be an amateur radio operator and the restricted frequency band 504 could be associated with one of the ~m~eur radio bands previously described with respect to FIG. 4.

The multicarrier modulation system does not utilize the frequencies in the restricted frequency band 504. Hence, as illustrated in FIG. 5, the frequencies within the restricted 25 frequency band 504 are not illustrated as carrying data as are the frequency tones 502 outside of the restricted frequency band 504. However, the presence of the radio interferer 506, even though within the restricted frequency band 504, has a d~ll hllelllal effect on the frequency tones outside the restricted frequency band 504 that are carrying data. Consequently, due to the radio interferer 506, the signals on the frequency tones 502 that are carrying data are corrupted by radio 30 interference. The amount of corruption will vary depending upon the tr~n~mitting power of the radio interferer 506 and how close the particular frequency of the tone is to the carrier frequency of the radio interferer 506.

.

CA 022~1946 1998-10-19 wo 97/40608 PCT/USg7/06384 In the example illustrated in FIG. 5, the radio intelrGi~r 506 transmits at a frequency that is contained within the restricted frequency band of the larger frequency range over which the multicarrier modulation system operates. The radio hlt~,lr~,rel could also be adjacent to the frequency range of the multicarrier modulation system. Still further, as discussed with reference to FIG. 11, the radio interferer could occur in the frequency range of the multicarrier modulation system but without regard to a restricted frequency band.

FM. 6 is a diagram 600 illustrating the amount of radio interference in~luce~l by the radio interferer 506 referenced in FIG. 6 on various frequency tones of a multicarrier modulation system. In this illustration, the height of the arrows on the frequency tones 602 indicate the o magnitude of the radio interference induced on that frequency tone by the radio i"t~,rt;l~l 506. As can be seen from FIG. 6, the m~gnit~lde of the radio interference in~l,nced on the frequency tones 602 decreases as the frequency becomes further removed from the carrier frequency of the radio interferer 506. In order to perform radio interference cancellation, the frequency tones outside of the restricted frequency band 504 need to be corrected for the radio interference. In other words, to cancel the radio interference, the radio inlt;lrer~nce induced on the frequency tones 602 outside of the restricted frequency band 504 needs to be estimated and then subtracted from the data received on the frequency tones 602. The number of the frequency tones that are removed in frequency from the carrier frequency of the radio interferer 506 which must be corrected (to mitigate the radio hltelrer~llce from the radio in~ rel~r 506 on those tones carrying data) depends 20 upon the processing techniques utilized and the degree of radio frequency mitigation desired.

FIG. 7 is a block diagram of a receiver 700 for a multicarrier modulation systemaccording to an embodiment of the invention. The receiver 700 receives radio signals 701 that have been transmitted by a multicarrier modulation system. I'he receiver 700 operates to process the received radio signals 701 to recover data that was transmitted by a transmitter of the 25 multicarrier modulation system. The tr~nsll,iller operates to transmit the data in blocks of data (e.g., DMT symbols). The cyclic prefix is added by the transmitter to provide a guard space to minimi7e inter-symbol interference and normally consists of a repetition of data from the end of a given data block.

The radio signals 701 are received by an analog radio frequency interference (RFI) 30 canceller 702. The analog RFI canceller 702 operates to mitigate radio intelre~llce in the analog domain, and then outputs radio frequency (RF) corrected radio signals 704. One suitable analog RFI canceller is described in International Patent. Application No. PCT/US97/ , filed April 17, 1997, entitled "Radio Frequency Noise Canceller", by Cioffi et al., which adaptively CA 022~1946 1998-10-19 ~o 97/40608 PCT/USg7/06384 estim~tes radio interference noise during data trAn~mi~ions using information obtained when no data is actually being tr~n.cmitt~.d The RF corrected radio signals 704 are supplied to an analog-to-digital converter 706. The correction to the radio signal 701 also ensures that the power level of the RF interference is below the saturation level for the analog-to-digital converter 706. The analog-to-digital converter 706 converts the RF corrected radio signals 704 to digital signals 708 which are output to a time domain equalization (TEQ) circuit 710. The time domain equalization circuit 710 produces time equalized digital signals 712. The time equalized digital signals 712 are then supplied to a cyclic prefix removal and windowing processor 714. The cyclic prefix removal and windowing processor 714 produces modified digital signals 716 which are supplied to a I o multicarrier demodulator 718. The processing performed by the cyclic prefix removal and windowing processor 714 is described in detail below with reference to FIG. 12. In one embodiment, the multicarrier demodulator 718 may be a Fast Fourier Transforrn (l~'~'l').The TEQ
circuit 710 limits the inter-symbol inl~lrer~nce by reducing the length of the channel impulse response.

The mnlti~rrier demodulator 718 outputs digital frequency signals 720 to a digital RPI
canceller 722. Although the received radio signals 701 have already undergone RP interference cancellation by the analog RFI canceller 702, additional RF hltel ~erel-ce cancellation is often needed. For example, additional RF hllelre~cnce cancellation is particularly needed when a radio h~l~.r~ ,r (e.g., an ~m~te~r radio operator) is transmitting in a restricted frequency band within a frequency range of a multicarrier tr~n~mi~ion system tr~ncmi~.~ion or when AM radio bro~ cting is nearby. The digital RFI canceller 722 outputs RF corrected digital signals 724 to a frequency-domain equalizer (FEQ) circuit 726. The FEQ circuit 726 outputs received digital signals 728 from which the transmitted data are obtained. The FEQ circuit 726 operates on each carrier (subchannel) and adaptively adjusts for the attenuation and phase delay of each carrier.

Radio i"lel~le,lce is initially modeled as a mod~ t~l windowed sinusoid in the time domain. FIGs. 8A-8C are representative diagrams illustrating examples of modulated sinusoids used to model radio frequency (RF) interference. The modulation of the sinusoid can take many forms as illustrated in FIGs. 8A-8C. In particular, in FIG. 8A, a time domain model modulates a sinusoid 800 using a rectangular envelope 802. In FIG. 8B, the time domain model modulates a sinusoid 804 with a linearly-varying envelope 806. In FIG. 8C, the time domain model modulates a sinusoid 808 with a (second-order) quadratically-modulated envelope 810. In general, the modulated sinusoid is modulated by an nth order polynomial modulation envelope.

CA 022~1946 1998-10-19 W O 97/40608 PCT~US97/06384 According to one aspect of the invention, the frequency domain model for RF interference that is utilized is derived and verified by the following ~li.ccu~ion For this di~c~l.c.cion, the time domain model illustrated in FIG. 8B is used as the exemplary embodiment. The RF interference is modeled in the time domain as a sinusoid multiplied by a linearly-mod~ te~ rectangular 5 window. More precisely, Equation (1) which follows provides the time domain model.
RFI(t) = rect(t)(l + at)cos[2~(fOt) + ~]
where rect(t) is a rectangular window,fO is a carrier frequency of the radio interference, a is a small positive constant, and ~ is a phase offset. This time domain model is equivalent to fitting a first-order polynomial to the modulation envelope of the RF interference within the time duration o of a data block (e.g., DMT symbol). The time domain model is suitable so long as the bandwidth of the radio interference (i.e., radio interferer) is much less than the symbol rate of the transmission system. For example, in the case of a amateur radio operator as the radio interferer and DMT as the transmission system, the bandwidth of the radio interferer is about 2.4 MHz which is substantially less than the symbol rate of the transmission system which is about 40 I S MHz.

Next, this time domain model is converted into the frequency domain for RF interference cancellation in the frequency domain. A Fourier Transform of the time domain model is performed to achieve the conversion. Equation (2) which follows details the conversion to the frequency domain.
, ~
F{rect(t)(l +at)} = ~ ~f ~ ~ , (2) The Fourier Transforrn of the cosine function of Equation ( I ) is a Dirac delta function at +f and -f . The negative frequency delta function is ignored because its contribution at the positive 25 frequencies is minim~l, particularly when non-rectangular windowing as discussed below is used.
However, the positive component could be used if the data transmission system does not utilize non-rectangular windowing.

As illustrated in Equation (2), there are two terms that drop off as 1 /f and one term that drops off as l/f2. LetfO = n + ~, where n is a frequency tone number, and o (0 < ~ < 1) being an - 30 offset amount of the carrier frequency of the RF hll~,relcnce from the frequency tone n.

CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 The resulting frequency domain model is as defined in Equation (3) which follows.

~RFln+m = + 2 . . . (3) m--~i (m--~i) where RFln+m is the RF h~ relt;nce to frequency tone m due to RF interference at a frequency n +
~, where A and B are complex numbers that must be determined for each symbol.
Further, when non-rectangular windowing is also used with the frequency domain model, the effect of the windowing can be approximated by multiplication by a single complex number Wm for each value of m, where Wm represents the phase rotation and additional attenuation (over that of rectangular windowing) due to the non-rectangular windowing operation. The complex number Wm is determined from the following Equation (4).

F{win(t)} ~=m F{win(t)}¦f=m (4) m F{rect(t)} f=ln sinc(m) where win(t) is the effective window used. Therefore, the resulting frequency domain model from Equation (3) now with non-rectangular windowing becomes as shown in Equation (5).

n+m m--~ (m _ ~ j)2 m where RFIn+n~ is the RF h~L~.re.~,lce to frequency tone m due to RF interference at a frequency n +
~, where A and B are complex numbers. Note that the frequency domain model requires that no data be transmitted on the frequency tones to either side of the frequency of the carrier frequency of the RF interference, namely frequency tones n and n+1, because these tones are used to determine values for A and B and~.
Instead of using three frequency tones to precisely determine A and B and~, the offset amount o can be approximated by the following Equation (6). Equation (6) is precise when the RF interference is a pure sinusoid.
Re{--} +¦Im{--}
Wl Wl ~i= . . . (6) ~Xn + 1~ ~Xn + 1~ R ~ Xn ~ + Xl ~
Rel w J+lml wl J lwoJ Im- woJ

where Xj represents the samples values for the frequency domain tones. The offset amount ~ is thus approximately equal to IXn+ll / {IXnl+lXn+ll~, which is accurate enough for çstim~ting RF
25 interference from an amateur radio operator. The frequency domain model has shown to be rather insensitive to small errors in the offset amount o.

CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 Then, using Equation (5) for tones n and n+l, two equations (Equations 7 and 8) can be written.
Xn A B
WO ~ j2 ~ ~ ~ (7) Xn+l A B
W' 1 - S + (I _ ~Sj)2 . . (8) - Simultaneously solving these two equations provides a technique to determine the complex parameters A and B of the frequency domain model. The complex parameters A and B are thus determined by the following equation.

A ~ 2 Xn B ~ )2 Xn + I ( ) Wl o The complex parameters A and B are determined, at each symbol, for each RF interferer, the Wm is a function of the windowing and varies with each of the frequency tones, and the offset amount o is computed once per symbol for each RF interferer being modeled. More generally, as noted above, o, A and B could be detel.l.ined by simultaneously solving three equations obtained from Equation (5) for three different tones (e.g., n, n+l and n+2), provided data is not transmitted on 5 these tones. Alternatively, the system could determine o as given by Equation (6) when the RF
interference is first detected, and then again use Equation (6) to average over many symbols to provide an estimate that becomes more accurate as the number of symbols averaged over increases.
In one embodiment, the frequency domain model provides sufficiently accurate modeling 20 of the RF interference that only the model parameter A, as computed in Equation (9), is used for cancellation, while the model parameter B is ~csume~ to be zero. With this simplification to the frequency domain model, the complexity is reduced, yet the frequency domain model still provides sufficient accuracy in modeling the RF interference in many cases. As an example, for RF interference caused by amateur radio operators, this simplification has shown to still provide 2s sufficiently accurate modeling (such as in a VDSL system). In other cases, the simplification may not be ~plopliate and the model parameter B should also be ~ltili7e(1, such as with higher bandwidth signals like AM radio signals.
Furthermore, higher order models could be likewise used to provide an even more accurate model for the RF i"~elrel~nce. However, the higher the order of the models used, the 30 greater the processing requirements to compute the parameters for the model. Hence, more CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 generally, the frequency domain model of Equation (3) according to the invention is in accordance with the following equation:
MO+I A
RFln+m ~ (m--~) . . (10) where RFI",m is the RF interference at a frequency tone m due to a radio interferer at frequency n, 5 o is an offset amount, MO is a model order for the frequency domain model, and { Ak ~ are complex numbers that are detelJ"i"ed at each symbol for each interferer. Hence, the frequency domain model derived above and defined by Equation (3) is a first order model (MO=I). Of course, when non-rectangular windowing is also used with the frequency domain model, the effect of the windowing can be approximated by multiplication by a single complex number Wm lO for each value of m, as was done in Equation (5).

FIG. 9 is a diagram of basic radio frequency (RF) cancellation processing 900 according to a basic embodiment of the invention. The RF cancellation processing 900 is preferably performed by a receiver or receiver portion of a transceiver of a multicarrier modulation system.

The RF cancellation processing 900 initially receives 902 frequency domain data. The 5 frequency domain data is data that has been transmitted by a transmitter of the multicarrier modulation system over a tr~n.~mission media to a receiver. Next, a restricted frequency band having radio frequency (RF) interference is identified 904. Then, assuming that a restricted frequency band has been identified as cont~ining RF interference, the frequency of the RF
interference within the restricted frequency band is estim~t~cl 906. After estimating the frequency 20 for the RF interference, the RF interference is estim~te(l 908 in accordance with the estim~t~d frequency and a frequency domain model for the RF h,te,~,~;nce. Thereafter, the estim~ted RF
interference is removed 910 from the frequency domain data. Following block 910, the RF
cancellation processing 900 is complete and ends.

FIGs. lOA and lOB are flow diagrams of digital RF cancellation processing 1000 25 according to an embodiment of the invention. It should be noted that the digital RF cancellation proce~.cing 1000 is associated with the processing performed by a receiver or receiver portion of a transceiver of a multicarrier modulation system upon receiving each symbol of a multicarrier tr;~nsmission system.

The digital RF cancellation processing 1000 initially receives 1002 data vectors X; for a 30 symbol. The data vectors X; are typically complex numbers for each of the frequency tones within a symbol. For example, in a 256-carrier DMT system, a data point Xj would be received for each of 256 frequency tones.

CA 022~1946 1998-lo-19 ~VO 97/40608 PCTrUS97/06384 Next, a restricted frequency band for RF cancellation processing is selected 1004. When there are multiple restricted frequency bands within the tr~n~mi.c~ion frequency range of the multicarrier tr~n.cmi~ion system, the processing described below is repeated for each of the restricted frequency bands. In any event, one of the restricted frequency bands is selected for RF
5 cancellation processing in which RFint~-r~rence produced in the restricted frequency band is cancelled from the received data vectors Xj. The RF cancellation processing 100 is described assuming at most one RF interferer is present in each of the restricted frequency bands.

Within the restricted frequency band that has been selected 1004, the largest data vector IXjlL within the restricted frequency band is determined 1006. Next, a decision block 1008 o determines whether the largest data vector l~jlL within the restricted frequency band is greater than a threshold. The value of the threshold will vary with system design but is normally set to a level such that a data vector IXjl in the restricted frequency band that is about 20 dB above the noise floor will exceed the threshold. When the largest data vector IXjlL is greater than the threshold, then the processing for the selected restricted frequency band continues.

Next, a largest adjacent data vector IXjl, ,,, iS determined 1010. Then, data vectors Xn and X~ are selected 1012 from the largest data vector IXjlL and the largest ~ ent data vector IXjl, The value of n provides an indication of an estimated frequency of the RF interference within the restricted frequency band because the received data vectors for the frequencies within the restricted frequency band do not carry information. In effect, at this point in the digital RF cancellation processing 1000, the carrier frequency of the RFh~ rel~llceis generally estimated to be between frequencies associated with n and n+1.

Next, an offset amount o is determined 1014 from the selected data vectors Xn and Xn+l.
For example, the offset amount o can be determined with Equation (6) with W0 ~1 and W, pre-stored in memory. Then, for the frequency domain model for the RF interference that has been selected (e.g., Equation (3)), model parameters A and B are computed 1016. As an example, Equation (9) can be used to determine the model parameters A and B. Once o, A and B have been determined, the frequency domain model for the RF interference is completed and may be used to cancel the RF interference from the received data vectors.
- A frequency tone is selected 1018 to receive cancellation. As previously noted, a predetermined number of the frequency tones that are adjacent to the restricted frequency band having the RF interference are selected such that they may be processed to cancel out the RF
interference. Although the canceling could be performed on all the frequency tones, it is computationally advantageous to pelroll.l cancellation only on a predeterrnined number of CA 022~1946 1998-10-19 97/40608 PCT~US97/06384 a~ljare.nt frequency tones. In any event, the selection 1018 of the frequency tone to receive cancellation operates to select one of these adjacent frequency tones. Then, for the selected frequency tone, the RF intel~lellce is estimated 1020 using the frequency domain model. Next, the estim~ted RF interference is subtracted 1022 from the data vector for the selected frequency tone. The subtraction performs the cancellation as illustrated in the following equation:
Xn+m(cancelled) = Xn+m(uncancelled) - RFIn+m where RFIn+m is obtained from Equation (10).

A decision block 1024 then determines whether cancellation of the RF interference has been completed. The decision block 1024 determines whether all of the frequency tones adjacent 0 to the restricted frequency band having the RF interference that require cancellation (e.g., the predetermined number) have received the necess~ry cancellation processing. Hence, if the cancellation has not been completed for all of the frequency tones to receive cancellation, the digital RF cancellation processing 1000 operates to select 1026 another frequency tone to receive cancellation. Following block 1026, the digital RF cancellation processing 1000 returns to repeat 15 block 1020 and subsequent blocks for the newly selected frequency tone. Note that for the newly selected frequency tone, the RF interference is again estimated for this newly selected frequency tone.

On the other hand, when a decision block 1024 determines that the cancellation for the frequency tones has been completed, a decision block 1028 determines whether all of the 20 restricted frequency bands have been processed. When all of the restricted frequency bands have not been processed, the next restricted frequency band is selected 1030 for RF cancellation proc.es~ing Following block 1030, the digital RF cancellation processing 1000 returns to repeat block 1006 and subsequent blocks so as to cancel RF interference in other restricted frequency bands. Alternatively, when the decision block 1028 determines that all of the restricted frequency 2s bands have been processed, the digital RF cancellation processing 1000 is complete and ends.

Further, when the decision block 1008 determines that the largest data vector IX;IL does not exceed the threshold, then the processing for canceling RF interference within the particular restricted frequency band is bypassed, and therefore not performed. In this case, the decision block 1008 causes the digital RF cancellation processing 1000 to jump to the decision block 1028 30 and thus bypass blocks 1010 through 1026.

CA 022=,1946 1998-10-19 W O 97/40608 PCT~US97/06384 In one implementation of the digital RF cancellation pr~ces~ing 1000, for a VDSLsystem, the processing is implemented by a digital ASIC coupled to or integrated with random access memory (RAM) and read only memory (ROM). The predetell,li,.ed number of adjacent tones to receive RF interference cancellation is 31 tones on each side of the RP interferer 5 (neglecting tones n and n+l), though the RP il-t~lr~ ce on the tones within the restricted frequency band need not be cancelled. In the case where the model order (MO) is one and B is assumed equal to zero, the l/(m-~) term in the frequency domain model for the RF interference can be computed using a first order polynomial approximation to avoid having to perforrn time-con.~l-rning divide operations. The coefficients aO and a, for the polynomial approximation are lo stored in memory for each value of m (a set for 0<~,<0.5 and a set for 0.5<~<1) and can thus be rapidly retrieved. The complex number Wm is also preferably 24-bits and stored in RAM. The data vectors for the frequency tones undergoing RF interference cancellation are pl~felably frequency domain data samplcs output from a ~1. Each of the restricted frequency bands can have its own threshold value.

ts Preferably, the computations for estim~ting the RF interference can be performed as follows. The largest element in the frequency band, and the largest element to either side of the largest element, are Xn and Xn+l. Next, interrnediate values o~, and ,~' are computed as follows.
lx ~ I ~
= 2 n X
where l/W, is held in RAM, ;md where W0 ~ 1. Then, intermediate values a and b are computed as follows.
¦R~ ~n~¦ ilm{~}

h=a+!h~p~l+llm{
.~ 4 2s The scaling down by a factor of 2 i~ done to prevent overflow during addition. The numbers a and b are then shifted such thal 0.5 < 1; c 1. Newton's method with eight iterations (I=0:7) is then used to find ~ = a/b. Set ~0 = 0.5, and then ~,+~ b - a) The model pal~ elt;r A (as scaled by a factor of 2) is then determined by the following equation.

A/2 = _~2~ +(1- ~2~.

CA 022~1946 1998-10-19 wo 97/40608 Pcr/uss7lo6384 The estimate of the RF interference for tone m is computed by forming rl =~aO +a r2 = AWm and thus the estimate of the RF interference becomes RFIn+m = 2(rl)(r2)-The estimated RF interference is computed is then subtracted from the data for the predet~ ed number of adjacent tones of the symbol (e.g., m = -31:32).
FIG. 11 is a flow diagram of AM radio frequency (RF) cancellation processing 1100 5 according to an embodiment of the invention. AM radio transmissions also cause RF interference to radio tr~nsmi~sions by a multicarrier modulation system. Unlike RF interference due to amateur radio operators, the AM RF interference is typically steadily present as AM radio stations tend to transmit 24 hours a day. The AM RF cancellation processing 1100 is preferably performed by a receiver or receiver portion of a transceiver of a mllltir~rrier modulation system.
20 The modeling of the RF interference described above equally applies to AM RF interference. For example, a first order model for the frequency domain model (e.g., Equation (5)) is also suitable for modeling AM R~ interference at VDSL rates.

The AM RF cancellation processing 1100 initially identifies 1102 AM RF interference during an initialization period in which no data is being transmitted. Then, the frequency of the 25 AM RF interference is estimated 1104. For example, by measuring the data signals received during the initi~li7~tion period in which no data is being tr~n~mittPd (as often the case with multicarrier modulation systems), the magnitude of the AM RF interference measured at different frequencies is found. Then, in this example, the areas in which the magnitude is maximized in~ tes a general location of the carrier frequency for the AM RF i~ ,rerellce. Thereafter, in this 30 example, the system can average the determined carrier frequencies over a period of time (e.g., many data blocks) to accurately determine the carrier frequency for the AM RF interference. By averaging the results of IXn+ll / {IXnl+lXn+~l} (or using Equation (6)) during the initi~li7~tion period, the offset amount o is able to be accurately determined and thus identifies the carrier frequency for the AM RF interference. Once the carrier frequency for the AM RF hllelrelence is 35 estim:lt~cl 1104, the initializatiofi is complete for this portion the AM RF cancellation processing 1100. Generally, the AM cancellation assumes that larger AM interferers are not close together in the AM frequency band.

CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 Thereafter when data is subsequently being transmitted or received, the AM RF
cancellation proce~.sing 1100 further opeldles to cancel the AM RF i,lte.re,cnce from the data signals being received. In the case of data tr~n~mi.csion, the frequency tones :~ljaçent to the estimated frequency of the AM RF interference are disabled 1106 so that no data is ~ld~ le~
s thereon. Here, at least the two frequency tones ~ cent to the estim~ed frequency of the AM RF
interference are disabled 1106 because the RF model uses these tones in modeling the RF
illtelr~,ence.

The cancellation of the AM RF inl~lrel~nce by the AM RF cancellation proce~.~ing 1100 is then as follows. The AM RF interference is estim~Pd 1108 in accordance with the e~li"~t d o frequency and a frequency domain model for the AM RF inlclr~lcnce. Thelearlel, the estim~ted AM RF interference is removed 1110 from the frequency domain data. Following block 1110, the RF cancellation processing 900 is complete and ends.

Non-rectangular windowing is generally known to reduce sidelobe levels in multicarrier modulation systems. See, e.g., Spruyt, Reusens and Braet, "Performance of improved DMT
]5 transceiver for VDSL, Alcatel Telecom TlE1.4 Submission, April 22-25, 1996. The non-rectangular windowing described by Spruyt et al. extends beyond the boundary of a symbol into a cyclic prefix and a cyclic suffix of the symbol.

The frequency domain model ~liccu$sed above preferably uses extended, non-rectangular windowing to cause sidelobes to attenuate faster so that the RF interference affects less frequency 20 tones. The particular type of non-rectangular windowing used can vary. FIG. 12 describes a possibly preferred type of non-rectangular windowing that is also another aspect of the invention that is useful not only with the RF cancellation techniques described herein but also by itself for mitigating hller~l;er interference in general.

FIG. 12 is a flow diagram of prefix removal and windowing proce.c~ing 1200 according 2s to an embodiment of the invention. Here, the windowing preferably performed is non-rectangular, extended windowing. The non-rectangular windowing acts to cause the sidelobes of the frequency tones to attenuate faster than rect~ng~ r windowing. The extended windowing means that the window width extends beyond the data symbol itself into a cyclic prefix. The cyclic prefix normally consists of a repetition of data from the end of the corresponding data symbol.
30 The cyclic prefix is a guard band that provides a guard time to reduce the intersymbol interference caused because channel responses are not ideal. As one example, in VDSL, the data symbol might have 512 samples and 40 samples of cyclic prefix. The prefix removal and windowing CA 022~1946 1998-10-19 W O 97/40608 rCTrUS97/06384 processing 1200 is preferably perforrned by the cyclic prefix removal and windowing processor 714 i}lustrated in FIG. 7.

The prefix removal and windowing processing 1200 initially receives 1202 X-samples of a DMT symbol and Y-samples of its cyclic prefix. For example, 512 samples of a DMT symbol and 40 samples of a cyclic prefix may make up a DMT symbol. FIG. 13 is a diagram illustrating a 512 sample DMT symbol 1300 (samples 40-551) with a 40 sample prefix 1302 (samples 0-39), and a non-rectangular, extended window. In FIG. 13, the non-rectangular, extended windowing extends from sample 20 to sample 551, with samples 20-39 being that portion that extends into the cyclic prefix. The processing of the X-samples of the DMT symbol and the Y-lo samples of the cyclic prefix are processed as follows.

An initial portion of the Y-samples of the cyclic prefix are dropped 1204 because they are no longer needed. A remaining ponion of the Y-samples are retained 1206 for later retrieval. The size of the rem~ining portion of the Y-samples depends on the amount of extended windowing being used. For example, with a 40 sample cyclic prefix, the size of the rem~ining portion of the 15 40-samples could by any whole number between 0 and 40. Next, a first portion of the X-samples of the DMT symbol are retained 1208. Then, a second portion of the X-samples of the DMT
symbol are modified 1210 in accordance with the retained samples of the remaining portion of the cyclic prefix and predeterrnincd multiplication coefficients. Following block 1210, the prefix removing and windowing processing 1200is complete and ends.

According to the prefix rcmoval and windowing processing 1200, the DMT symbol and its prefix have been processed such that the resulting samples have been filtered such that an initial group of samples of the pre~lx ~ n~c~ved and then extended non-rectangular windowing processing is performed on lhc rcmaining samples. The extended non-rectangular windowing operates to multiply the samplc~ Or thc remaining portion of the cyclic prefix by a raised-cosine-2s function (or other smoothing ~unclion) representing the non-rectangular portion of the window, and then combines the resultinF ~aluc into the samples of the second portion of the X-samples.
The advantage of the extended non-rect~ngular windowing operation is that the effective sidelobe levels data attenuate faster which i~ generally advantageous in a multicarrier modulation system.
In the case where the extended non-rcctangular windowing is used with the RF cancellation 30 techniques according to the invention, the advantage of the extended non-rectangular windowing is that RF cancellation can be performed on fewer adjacent frequency tones which reduces processing needed to compensate for the RF interference. The saved processing time which can be important in high-speed systems such as multicarrier modulation systems (e.g., VDSL). The CA 022~1946 1998-10-19 W O 97/40608 PCTrUS97/06384 extended non-rectangular windowing according to the invention further reduces the colllpu~alional burden required to implement the exten~ed, non-rectangular windowing. The following examples helps to explain the additional computational savings offered by this aspect of the invention.

An example of the prefix removal and windowing processing 1200 is explain for a case where 32-sample extended windowing is utilized with 512 DMT frequency tones and 40-samples of cyclic prefix. The values xO through x55l represent a single DMT symbol with its cyclic prefix, and the values wO through w3l are window taps that are preferably stored in RAM. In this example, the prefix removal and windowing processing 1200 is as follows:
lo Discard xO through X7 Store x; for i = 8 to 39 xj=xjfori=40toS19 Form Xs20+i= X52~j+(X8+j-X52~j)Wj~ for i = 0 to 31.
Note that x52~, j = ( 1 -wj)x520+j + wjx8+j = x52~ + (x8+j - x520~ j )wi, and the implementation requires 32 real multiply operations and 64 addition operations per DMT symbol. In contrast, the conventional approach would utilize 64 real multiply operations and 32 or 64 addition operations per DMT symbol. Given that the computational burden to perform a multiply operation is significantly greater than the computational burden for an addition operation, the ability of the invention to save 32 multiply operations is noteworthy.
It should be understood that the present invention may be embodied in many forms and modulation schemes (e.g., Discrete Wavelet Multi-tone Modulation (DWMT)) at both the central and remote station locations without departing from the spirit or scope of the invention. For instance, although the specification has primarily described the invention in the context of subscriber line based high speed data tr~n.cmi.csion systems, the invention may be used in other 2s systems which experience signific~nt narrow band interference or have restricted frequency bands of RF interference within their designated tr~n.cmi.c.cion bands.
The many features and advantages of the present invention are a~ar~nt from the written description, and thus, it is intended by the appended claims to cover all such features and advantages of the invention. Further, since numerous modifications and changes will readily occur to those skilled in the art, it is not desired to limit the invention to the exact construction and operation as illustrated and described. Hence, all suitable modifications and equivalents may be resorted to as falling within the scope of the invention.
..

Claims (52)

1. A method for mitigating radio frequency (RF) interference in a multicarrier modulation system, said method comprising the operations of:
(a) obtaining frequency domain data associated with a frequency band;
(b) identifying a restricted frequency sub-band within the frequency band;
(c) estimating a frequency of the RF interference within the restricted frequency sub-band;
(d) estimating the RF interference in accordance with a frequency domain model for the RF interference and the estimated frequency of the RF interference; and (e) removing the estimated RF interference from the frequency domain data.
2. A method as recited in claim 1, wherein the frequency domain data is provided in a plurality of frequency tones within the frequency band, and wherein the frequency domain model is produced in accordance with the following equation:
where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.), .delta. is an offset amount, MO is a model order for the frequency domain model, and A k is a complex number that is determined for each frequency tone m.
3. A method as recited in claim l, wherein the RF interference is due to radio transmissions by an amateur radio operator.
4. A method as recited in claim 1, wherein the frequency restricted sub-band is approximately one of: 1.8 to 2.0MH z;3.5 to 4.0MH z;7.0 to 7.3MH z and 10.1 to 10.15MH z.
5. A method as recited in claim 1, wherein the frequency domain data contains a plurality of frequency domain data samples, and wherein said estimating of the frequency of the RF interference comprises the operations of:
determining a largest data sample of the frequency domain data samples within the restricted frequency sub-band, and determining a largest adjacent data sample that is adjacent to the largest data sample; and determining the frequency of the RF interference within the restricted frequencysub-band based on the largest data sample and the largest adjacent data sample.
6. A method as recited in claim 1, wherein the frequency domain model is based on a time domain model for the RF interference in which the RF; interference is modeled as a windowed, modulated sinusoid.
7. A method as recited in claim 6, wherein the sinusoid is modulated by a windowed, modulation envelope.
8. A method as recited in claim 6, wherein the sinusoid is modulated by a linearly-varying, windowed, modulation envelope.
9. A method as recited in claim 6, wherein the sinusoid is modulated by a n th order polynomial modulation envelope.
10. A method as recited in claim 1, wherein the frequency domain data contains a plurality of frequency domain data samples, wherein said estimating the RF interference estimates the RF interference for at least a portion of the frequency domain data samples, and wherein said removing of the estimated RF interference from the frequency domain data comprises, for each of the frequency domain data samples in the portion, the operation of subtracting from the frequency domain data sample the estimated RF interference on that frequency domain data sample.
I l . A method as recited in claim 10, wherein the frequency domain data contains a plurality of frequency domain data samples, and wherein said estimating of the frequency of the RF interference comprises the operations of:
determining a largest data sample of the frequency domain data samples within the restricted frequency sub-band, and determining a largest adjacent data sample that is adjacent to the largest data sample; and determining the frequency of the RF interference within the restricted frequencysub-band based on the largest data sample and the largest adjacent data sample.
12. A method as recited in claim 11, wherein the frequency domain model is based on a time domain model for the RF interference in which the RF interference is modeled as a modulated sinusoid.
13. A method as recited in claim 12, wherein the RF interference is due to radio transmissions by an amateur radio operator.
14. A method as recited in claim 13, wherein the frequency domain data is provided in a plurality of frequency tones within the frequency band, and wherein the frequency domain model is produced in accordance with the following equation:
where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.), .delta. is an offset amount, W m is an attenuation factor due to time domain windowing and varies with each of the frequency tones, and A and B are complex numbers.
15. A method as recited in claim 14, wherein A and B are model parameters and are determined by the following equation:

where the complex parameters A and B are determined once for each symbol, and the offset amount o is computed once per symbol for each RF interferer being modeled.
16. A method as recited in claim 1, wherein the frequency domain data contains a plurality of frequency domain data samples, wherein said method further comprises the operation of comparing the frequency domain data samples within the restricted frequency band with a threshold amount, and wherein, for the restricted frequency band, at least one of said estimating (d) and said removing (e) are bypassed when said comparing determines that the frequency domain data samples are less than the threshold amount.
17. A method as recited in claim 1, wherein no data is transmitted in the restricted frequency sub-band.
18. A method as recited in claim 1, wherein said obtaining (a) of the frequency domain data is initially received as time domain data, the time domain data undergoes a time domain windowing operation, and thereafter the windowed time domain data is converted to the frequency domain.
19. A method for mitigating radio frequency interference in a multicarrier modulation system, said comprising the operations of:
prior to data transmission, identifying AM radio interference in the multicarrier modulation system;
estimating a frequency of the AM radio interference;

disabling certain frequency tones of the multicarrier modulation system ~ cent to the estimated frequency of the AM radio interference from carrying frequency domain data during the data transmission;
thereafter, during or following data reception, estimating the AM radio interference in accordance with a frequency domain model for the AM radio interference and the estimated frequency of the AM radio interference;
and removing the estimated AM radio interference from the frequency domain data.
20. A method as recited in claim 19, wherein said identifying of the AM radio interference is performed during an initialization period of the multicarrier modulation system that occurs prior to data transmission.
21. A method as recited in claim 19, wherein the frequency domain data contains a plurality of frequency domain data samples, and wherein the frequency domain data is initially received as time domain data, the time domain data undergoes a time domain windowing operation, and thereafter the windowed time domain data is converted to the frequency domain.
22. A method as recited in claim 19, wherein the AM radio interference resides within a AM
radio band, wherein the frequency domain data contains a plurality of frequency domain data samples, and wherein said estimating of the frequency of the AM radio interference comprises the operations of:
determining a largest data sample of the frequency domain data samples within a frequency range, and determining a largest adjacent data sample that is adjacent to the largest data sample; and determining the frequency of the AM radio interference within the frequency range based on the largest data sample and the largest adjacent data sample in a portion of the radio band.
23. A method as recited in claim 22, wherein the frequency domain model is based on a time domain model for the RF interference in which the RF interference is modeled as a windowed, modulated sinusoid.
24. A method as recited in claim 23, wherein the sinusoid is modulated by a windowed, modulation envelope.
25. A method as recited in claim 23, wherein the modulated is modulated by a linearly-varying, windowed, modulation envelope.
26. A method as recited in claim 23, wherein the sinusoid is modulated by an n th order polynomial modulation envelope.
27. A method as recited in claim 19, wherein the frequency domain data contains a plurality of frequency domain data samples, wherein said estimating the AM radio interference estimates the AM radio interference for at least a portion of the frequency domain data samples, and wherein said removing of the estimated AM radio interference from the frequency domain data comprises, for each of the frequency domain data samples in the portion, the operation of subtracting from the frequency domain data sample the estimated AM radio interference on that frequency domain data sample.
28. A method as recited in claim 27, wherein the AM radio interference resides within a AM
radio band, wherein the frequency domain data contains a plurality of frequency domain data samples, and wherein said estimating of the frequency of the AM radio interference comprises the operations of:
determining first and second largest data samples of the frequency domain data samples within the portion of the frequency domain data samples; and determining the frequency of the AM radio interference based on the first and second largest data samples in a portion of the radio band.
29. A method as recited in claim 28, wherein the frequency domain model is based on a time domain model for the AM radio interference in which the AM radio interference is modeled as a modulated sinusoid.
30. A method as recited in claim 29, wherein the AM radio interference is due to radio broadcasts by radio stations.
31. A method as recited in claim 30, wherein the frequency domain data is provided in a plurality of frequency tones, and wherein the frequency domain model is produced in accordance with the following equation:

where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.) .delta. is an offset amount, W m is an attenuation factor due to time domain windowing and varies with each of the frequency tones, and A and B are complex numbers.
32. A method as recited in claim 31, wherein A and B are model parameters and are determined by the following equation:

where the complex parameters A and B are determined once for each symbol, and the offset amount o is computed once per symbol for each RF interferer being modeled.
33. A method as recited in claim 19, wherein the frequency domain data contains a plurality of frequency domain data samples, wherein said method further comprises the operation of comparing the frequency domain data samples with a threshold amount, and wherein at least one of said estimating the AM radio interference and said removing of the estimated AM radio interference are bypassed when said comparing determines that the frequency domain data samples are less than the threshold amount.
34. A method as recited in claim 19, wherein said estimating of the AM radio interference further being in accordance with the frequency domain data on the certain frequency tones on which no data, just AM radio interference, is present.
35. A method as recited in claim 19, wherein said estimating of the frequency of the AM
radio interference is performed while data is not being transmitted.
36. A method as recited in claim 19, wherein the frequency domain data is provided in a plurality of frequency tones, and wherein the frequency domain model is produced in accordance with the following equation:

where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.), .delta. is an offset amount, MO is a model order for the frequency domain model, and A k is a complex number.
37. A method for digitally filtering multicarrier modulation samples to reduce sidelobe interference from a radio frequency (RF) interferer, the multicarrier modulation samples occur at predetermined frequency tones and form a multicarrier modulation symbol, said method comprising the operations of:
receiving x samples of a multicarrier modulation symbol and y samples of a cyclic prefix associated with the multicarrier modulation symbol, the y samples of the cyclic prefix preceding the x samples of the multicarrier modulation symbol;
discarding an initial portion of the y samples of the cyclic prefix associated with the multicarrier modulation symbol;
storing a remaining portion of the y samples of the cyclic prefix associated with the multicarrier modulation symbol;
retaining a first portion of the x samples of the multicarrier modulation symbol without modification; and modifying a second portion of the x samples of the multicarrier modulation symbol in accordance with the stored samples of the remaining portion of the y samples of the cyclic prefix and predetermined multiplication coefficients.
38. A method as recited in claim 37, wherein said receiving of the x samples of a multicarrier modulation symbol and y samples of a cyclic prefix associated with the multicarrier modulation symbol is a stream of data received over a transmission media from a transmitter of a multicarrier modulation system.
39. A method as recited in claim 38, wherein the transmission media is a subscriber line.
40. A method as recited in claim 37, wherein for each x samples of the multicarrier modulation symbol, said method uses j multiply operations and 2j addition operations for performing said modifying, where j is an integer representing the number of samples in the remaining portion of the y samples of the cyclic prefix.
41. A method as recited in claims 40, wherein the predetermined multiplication coefficients are associated with a raised cosine function.
42. A method as recited in claim 37, wherein said modifying of the second portion of the x samples of the multicarrier modulation symbol comprises:
retrieving an appropriate one of the predetermined multiplication coefficients;

determining a difference amount between corresponding pair of samples of the remaining portion of the y samples of the cyclic prefix and the second portion of the x samples of the multicarrier modulation system;
multiplying the difference amount with the appropriate one of the predetermined multiplication coefficients to produce an adjustment amount; and adding the adjustment amount to the sample of the second portion of the x samples of the corresponding pair.
43. A method for digitally filtering DMT samples to reduce sidelobe interference from a radio frequency (RF) interferer to frequency tones of a DMT symbol, said method comprising:
receiving X samples of a DMT symbol and Y samples of a cyclic prefix associated with the DMT symbol;
discarding an initial portion of the Y samples of the cyclic prefix;
storing a remaining portion of the Y samples of the cyclic prefix;
retaining a first portion of the X samples of the DMT symbol without modification; and modifying a second portion of the X samples of the DMT symbol in accordance with the stored samples of the remaining portion of the Y samples of the cyclic prefix and predetermined multiplication coefficients.
44. A method as recited in claim 43; wherein said modifying operates to attenuate sidelobe interference from a radio frequency (RF) interferer at a rate faster than would occur without said modifying.
45. A method as recited in claim 43, wherein said method reduces the number of the frequency tones of the DMT symbol that are closest to the frequency of the RF interferer than are seriously impacted by the RF interference.
46. A receiver for a multicarrier modulation system, comprising:
an analog-to-digital (A/D) converter, said A/D converter receives analog signals that have been transmitted to said receiver over a transmission media and converts the analog signals to digital time domain signals;
a multicarrier demodulator operatively connected to said A/D converter, said multicarrier modulator receives the digital time domain signals and converts the digital time domain signals into digital frequency domain data; and a digital RF interference canceller operatively coupled to said multicarrier demodulator, said digital RF interference canceller mitigates the effect of RF interference on the digital frequency domain data by modeling the RF interference in accordance with a frequency domain model.
47. A receiver as recited in claim 46, wherein said digital RF interference canceller mitigates the effect of RF interference on the digital frequency domain data by estimating a frequency of the RF interference, estimating the RF interference in accordance with the frequency domain model for the RF interference and the estimated frequency of the RF interference, and removing the estimated RF interference from the digital frequency domain data.
48. A receiver as recited in claim 46, wherein the digital frequency domain data is provided on a plurality of frequency tones used by the multicarrier modulation system, and wherein the frequency domain model is produced in accordance with the following equation:
where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.), .delta. is an offset amount, MO is a model order for the frequency domain model, and A k is a complex number.
49. A receiver as recited in claim 46, wherein the digital time domain signals include a plurality of multicarrier modulation symbols carrying data, each of the symbols having a cyclic prefix, wherein said receiver further comprises:
a cyclic prefix removal and windowing processor operatively connected between said A/D converter and said multicarrier demodulator, said processor performs a time domain windowing operation on the symbols, the time domain windowing includes, for each symbol, adding a portion of the cyclic prefix multiplied by a predetermined coefficient to a rear portion of the symbol.
50. A receiver as recited in claim 49, wherein the digital frequency domain data is provided on a plurality of frequency tones used by the multicarrier modulation system, and wherein the frequency domain model is produced in accordance with the following equation:
where RFI n+m is the RF interference at a frequency tone n+m due to a radio interferer at frequency (n+.delta.), .delta. is an offset amount, A k is a complex number, MO is a model order for the frequency domain model, and W m is an attenuation factor associated with the time domain windowing operation.
51. A receiver as recited in claim 49, wherein said receiver further comprises:

an analog RF canceller operatively connected to reduce RF interference from the analog signals prior to their being supplied to said A/D converter.
52. A receiver as recited in claim 49, wherein the time domain windowing is extended windowing, wherein, for each symbol, the window extends beyond the boundaries of the symbol into the cyclic prefix.
CA002251946A 1996-04-19 1997-04-17 Digital radio frequency interference canceller Abandoned CA2251946A1 (en)

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US1625296P 1996-04-19 1996-04-19
US1625196P 1996-04-19 1996-04-19
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US60/016,252 1996-04-19
US08/834,503 1997-04-04
US08/834,503 US6014412A (en) 1996-04-19 1997-04-04 Digital radio frequency interference canceller

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US6404830B2 (en) 2002-06-11
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