CA1184256A - Direct modulation fm demodulator - Google Patents
Direct modulation fm demodulatorInfo
- Publication number
- CA1184256A CA1184256A CA000396534A CA396534A CA1184256A CA 1184256 A CA1184256 A CA 1184256A CA 000396534 A CA000396534 A CA 000396534A CA 396534 A CA396534 A CA 396534A CA 1184256 A CA1184256 A CA 1184256A
- Authority
- CA
- Canada
- Prior art keywords
- signal
- mixers
- output
- input
- phase
- Prior art date
- Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
- Expired
Links
Classifications
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
- H03D3/007—Demodulation of angle-, frequency- or phase- modulated oscillations by converting the oscillations into two quadrature related signals
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D3/00—Demodulation of angle-, frequency- or phase- modulated oscillations
- H03D3/02—Demodulation of angle-, frequency- or phase- modulated oscillations by detecting phase difference between two signals obtained from input signal
- H03D3/22—Demodulation of angle-, frequency- or phase- modulated oscillations by detecting phase difference between two signals obtained from input signal by means of active elements with more than two electrodes to which two signals are applied derived from the signal to be demodulated and having a phase difference related to the frequency deviation, e.g. phase detector
-
- H—ELECTRICITY
- H03—ELECTRONIC CIRCUITRY
- H03D—DEMODULATION OR TRANSFERENCE OF MODULATION FROM ONE CARRIER TO ANOTHER
- H03D7/00—Transference of modulation from one carrier to another, e.g. frequency-changing
- H03D7/16—Multiple-frequency-changing
- H03D7/165—Multiple-frequency-changing at least two frequency changers being located in different paths, e.g. in two paths with carriers in quadrature
Abstract
ABSTRACT:
A direct modulation FM demodulator in which an incoming FM signal (2a sin(.omega. + .delta. .omega.)t) is demodulated to baseband (.delta. .omega.)by applying it to first and second mixers. The output (b sin .omega. t) of a local oscillator is applied to one of the mixers and is also applied to the other of the mixers via a 90° phase shifter. The phase of the output of one of the mixers is shifted so as to be in phase or in antiphase to the other and is applied together with the output of the other of the mixers to a divider, the output of which comprises the baseband signal which is substantially independent of fading. The channel phase shifting may be done using a differentiator or an integrator.
A direct modulation FM demodulator in which an incoming FM signal (2a sin(.omega. + .delta. .omega.)t) is demodulated to baseband (.delta. .omega.)by applying it to first and second mixers. The output (b sin .omega. t) of a local oscillator is applied to one of the mixers and is also applied to the other of the mixers via a 90° phase shifter. The phase of the output of one of the mixers is shifted so as to be in phase or in antiphase to the other and is applied together with the output of the other of the mixers to a divider, the output of which comprises the baseband signal which is substantially independent of fading. The channel phase shifting may be done using a differentiator or an integrator.
Description
5~
4.9~81 1 PHB 3~7~o "A DIRECT MODULATION FM DEMODULATOR"
- The present invention relates to a direct modulation (frequency modulatioIl) demodulator. More particularly, the present invention relates to an F~I demodulator in which the use of inductors for filtering are avoided and the output signal is substantially independent of fading which is particularly useful in mobile receiver applications in which the strength of an input signal c&n vary quite rapidlyO
It is known to demodulate frequency-modulated signals by a method which involves sampling in some way. However, a disadvantage of this method is that when the signal has been mixed down to audio frequenciesg the s&mpling produces harmonics which lie in the audio band and cause distortion.
There also exist a number of proposals for demodulating FM
signals which do not involve sampling. British Patent Specification 1,530,602 discloses a demod~Llator comprising first and second parallel arranged signal paths whose inputs are connected to receive an incorning FM signal and whose outputs are connected to a difference amplifier. Each of the first and second si~nal paths comprises a first mixer having output connected to a low pass filter whose output is in turn connected via an automatic level control (ALC) amplifier to one input of a second mixer the output of which is connected to a respective input of the difference &mplifier~ A first signal input to the first mixer in each pa-th is the incoming FM
signal. A second signal input of the first mixer in the second path is a reference signal and the second signal input of the first mixer in the first path is the same reference signal which has been phase shifted by 9O. The outputs of the fir~t mixers are the sum and difference components of mixing. The low pass filter in each path passes the difference components, namely a cos ~ ~ t in the first path and a sin ~ ~ t in the second path, where 2a is the ampli-tude .~, 4.9~81 2 PI~B 32760 of the input signal, ~ ~ i6 the angular frequency differenee component and the amplitude of a signal from a loeal oseillator is unity. The ALC amplifiers enable the signals processed thereafter to have a st~ldard level of signal with which to work and enable the receiver to have a large working ranoee signal.
The output signals from the ALC amplifiers are supplied to a first signal input of the second mixer in its respective signal path and to respec-tive differentiators which differentiate their respective signals to give terms with arnplitude/frequeney slope and a quadrature phase shift, the output of the differentiator whose input is connected to the first path i8 connected as a second signal input to the second mixer in the second path, conversely the output of the other differentiator is connected as a second signal input to the second mixer of the first path.
The signals applied to the inputs of the difference amplifier comprise a2~ ~ cos2 S ~ t from the first path and -a2 ~ ~ sin ~ ~ t from the second path and the output from the difference &mplifier is a ~ ~ . Since this output comprises a power term, namely a2, then the output is dependent on the amplitude of the input signal which may vary considerably and rapidly if the demodulator is part of a mobile receiver.
Also known is a sine cosine frequency traeker which is disclosed in British Patent Specification 1,363,396. The frequency tracker includes a frequency discriminator which is very similar to the demodulator of British Patent Specification 1,530,602 but in which there are no ALC amplifiers. Iihe output from a difference amplifier is A2 ~ where A is the amplitude of the input signal and ~ is the difference signal between the incoming signal and a reference derived from a sweep oscilla-torO In the case of the tracking system the A2 ~ term is required in order to provide a control signal for the sweep oseillator.
United States Patent Specification 3,s68,o67 also disclo~ies a frequency diseriminator whieh is very similar to that disclosed in British Patent Specification 1,363$396~ ~his too produces an - - - -4.9~8~ 3 PHB ~,2760 i _ output which is amplitude dependent.
An object of -the present invention is to provide an FM
modulator in which amplitude and sinusoidal variations are reduced or eliminated.
According to the present invention there is provided a direct modulation FM demodulator in which an input signal ~nd a local oscillator signal are mixed in two mixers to provide outputs which are in quadra-ture phase relationship with each other, a low frequency output of one of the two mixers is phase shifted by 90 more than the phase shift in a low frequency output of the other of the two mixers to render the two signals in phase or antiphase with each other and in which the two signals are applied to a divider to obtain as an output a modulating signal contained in the input signal.
~ he phase shifting of one of the first and second mixing operations may be accomplished by di ferentiating or integrating the signal from said one mixer or by using a pair of phase shifting networks, one in each channel if only the sign of S ~
is to be determined or resolved, for example as may be the case in data transmission.
An advantage of the demodulator in accordance with the present invention is that it enables demodula-tion to be carried out at baseband due to its continuity (that i9 because there is no sampling) and it is resistant to th0 ef~ects of fading.
! The quadrature mixing of the input signal and the local oscillator signal is enabled by shifting the phase of one or another of these sig~nals by 90 prior to its being applied to ona of the two mixers.
An embodiment of a demodulator made in accordance with the present invention comprises a signal input, first and second mixers each having first and second inputs, said first inputs being connected to the signal input1 means producing a first local oscillator signal which is supplied to the second input of -the first mixer, means produc;ng a second local oscillator , . . .
,, . _ .. . . _ . _ _ _ . . . . . . . ... . .
; 4.9.81 4 PH~ 32760 I
_ _ signal which is of the same frequency as, and shifted in pha6e -- ---- by 90 relative to, the first local oscillator signal and supplyin~ it to the second input of the second mixer7 a divider having first and second inputs and an output constituting a 6 si~nal output, the first input being cornected to receive a low frequency output of one of the first and second mixers1 and the second input being connected to receive a low frequency output of the other of the first and second mixers which has been shifted in phase by 90 relative to the phase of the signal applied to the first input~ thus making the applied signals in phase or in antiphase.
The divider conveniently comprises a multiplier in a feedback arrangement. In the case of an analogue multiplier means are provided for inverting the polarity of the input signal forming the denominator in order to keep it positive~
In an embodiment of the present invention a further low pass filter is connected in the signal path from each of the first and second mixers to remove adjacent channel :interference.
If desired automatic galn control means are provided in at least one signal path from one of the first and second mixers.
Fmbodiments of the present invention will now be described, by way of example, with reference to the accompanying drawings, wherein:
Figure 1 is a block schematic circuit diagram of one embodiment of the present invention7 Figure 2 is a block schematic circuit diagram of another embodiment of the present invention7 and Figure ~ is a schematic circuit diagram of an embodiment of the type shown in ~igure 1~
Referring to Figure 1 an input FM signal 2a sin ( ~ ~ o ~ )t is applied to a first signal input of rirst and second mixers 12, 14, respectively, via an input terminal 10~ ~ signal generator or lccal oscillator 16 produces a signal, b sin ~ t, having a frequency corresponding to the carrier wave of the input signal~
This local oscillator signal is applied to a second signal input . , , , _ . . _ . _ _ _ ., _ , _ . . . _ . , .. . . , . , _ . ., 4.9.~1 5 PHB 32760 of the second mixer 14, and is applied to a phase shifter 18 which . .
shifts the phase of this signal by 90 A The phase shifted si~nal is applied to a second input of thc first mixer 12. The mixers 12, 14 which may include low pass filters produce outputs at base band frequency, in the case of the first mixer 12 the output can be represented by ab sin ~ ~ t and in the case of the second mixer 14 the output can be represented by ab cos ~ ~ t Both these signals will contain amplitude and sinusoidal variations.
These variations can be reduced and/or eliminated using a divider 22. In order to do this the phase of one of the signals has to be shifted b~ 90 in order to render it in phase or antiphase with the other of the signals. In Figure 1 this is done by applying the output, ab sin ~ ~ t, of the first mixer 12 to a differentiation circuit 20, the output of which, ~ ~ . ab cos ~ ~ t, is applied to an input X of the divider 22. A second input ~
of the divider receives the output, ab cos ~ ~'t, from the second mixer 14. The divider carries out the operation X/Y (iOeO
. ab cos ~ ~ t/ab cos ~ ~ t) to give an output signal ~ ~ on the output Z of the divider 22. The output signal ~ ~J
corresponds to the modulation component of the input iignal applied to the terminal 10 c~nd is independent of ~nplitude a Consequently the effects of fading on the output signal are reduced.
The embodiment shown in Figure 2 differs from the embodiment of Figure 1, by the fact the connections from the mixers 12, 14 to the lnputs X, Y, of the divider 22 are different. The input X of the divider 22 comprises the output ab sin ~ t from the mixer 12. The output ab cos ~ ~ t o-f the mixer 14 is integrated in an integrator 24 to provide a ~0 phase shift having an output (1/ S ~ ).ab sin ~ ~ to This comprises the input Y to the divider.
The output Z of the divider 22 again comprises ~ ~ .
In the embodiments illustrated in Figrures 1 and 2 if the circuit components in the signal paths do not proride low pass filtering to select the difference products of mixing then separate low pass filters 26, 28, shown in broken lines, should be coupled to the outputs of the mi~ers 12, 14.
.... .. . .... . .. . _ . .. .
4 9, ~1 6 PHB 32760 Referring now to Fibure 3 which i5 a schematic circuit diabram of an embodiment of the type shown in Figure 1 which uses a differer.tiator. For the sake of consistency and ease of explanation the main elements of the circuit have been enclosed in broken lines and identified by the same reference numerals as used in Figure 1. Two low pass filters 26 and 28 are connected between the mixer 12 and the differentiator 20 and between the mixer 14 and the divider 22, respectively, These filters Z6, 28 serve to filter out the high frequency lD image component and other unwanted signals in the outputs of the mixers 12 and 14.
In the following description since the mixers 12~ 14 and low pass filters 26, 28 are identical only the mixer 12 and the filter 26 will be described in detail~ The mixer 12 comprises a multiplier type SL 640C integrated circuit 30, -the pin numbers of which are shown within the rectangle. The input terminal 10 is connected to the pin 7 by way of a capacitor 32 having a value 2n7 farads. The output of the phase shifter 18 is connected to a pin 3 by way of a capacitor 34 haring a ~ value of 4r.7 farads. The pins 3 and 7 are biased ~rom a +6 volt rail by means of 10KJ-J potentiometers 36, 38 and 330K ~
resistors 40, 42, respectively. The signal output of the mixer 12 is derived from the pin 5 of the integrated circuit 300 The other connections of the integrated circuit 30 are as sho~m and the ralue of the capacitor between pin 2 and ground is 10 uF.
The low pass filter 26 is based on an operational amplifier 44, for example amplifier type 531, having negative feedback via 27K J~ resistor 46. The relevant pins of the type 531 amplifier are shown within the triangular symbol representing the amplifier.
A capacitor of 47 pF is cornected between a pin 8 o~ the amplifier 44 and its output. The output, pin 6, of the mixer 12 is connected to the inverting input, pin 2, of the amplifier 44 via an R~C.
filter network comprising series connected resistors 48 (4K7~ and 50 (2K7) and a capacitor 52 (3n3F) connected between a junction of the resistors 48~ 50 and ground. The non-inrerting input, pin .. . .. .. . .. .. . .
4;9.81 ~ P~rB ~2760 3, of the ~nplifier is bia~ed using a potentiometer 54 (4K7) and a series resistor 56 (10K) connected between the 6 volt rail and ground .
The differentiator 20 is based on ano-ther operational amplifier 58, for example amplifier -type 531. As is customary there is a feedback resistor 60 (5K6) connected between the amplifier output, pin 6, and the inverting input~ pin 29 of the amplifier 58, which input is connected to the output of the filter 26 by way of a capacitor 62 (10 ~). A non-inverting input, pin 3, of the amplifier 58 is connected to ground~ A
capacitor of 220 pF is connected between pin 8 on the amplifier 58 and its output.
As the oscillator 16 is a standard crystal oscillator circuit based on an npn transistor 64 type BF 494 it will not be described further. The output of the oscillator is connected -to a phase splitting arrangement comprising an npn transistor 66~ type ~F
494. As such an arrangement is well-known it will not be described further. The non-inverted sign~l is tapped-off from the emitter circuit of the transistor 66 and is applied by way of a capacitor 68 (4n7) to pin 3 of the integrated circuit in the mixer 14. The inverted signal is derived from the collector of the transistor 66 and applied to the phase shifter 18 which ad~u~ts the phase o~
the signal by 90.
The phase shifter 18 comprises two npn transistors 70, 72, type BF 494, which are operated as emitter followersO The inverted signal from the phase splitting arrangement is applied to the base electrode of the trarsistor 70 by way of an R.C. phase balancin~
network comprising a series variable resistor 74 (470 f~) and a capacitor 76 (100 p~) coupled to ground. An R.C. filter 78 couples the emitter o~ the transistor 70 to the base of the transistoI 72~ The 90 phase shifted signal is derived from the tap of a potentiometer 80 in the emitter circuit of the tr~lsistor 72 and is applied via the capacitor 34 to pln 3 of the integrated circuit ~Or The potentiometer 80 provides amplitude balance to the derived signal~
.. . . , , ... _ . . , _ .. _ . . .... _ . . . _ . . . .
5~j 4.9.81 8 Pl-IB 327~0 ------- The output of the differentiator 20 comprises the signal -~~~-X which is applied a& one input to the divider 22 and the output of the low pass ilter 28 comprises the signal Y which is applied as another input to the divider 22. In the illustrated embodiment the divider 22 is based on an analogue divider 82, for example an Analog Devices AD 53~D, the pin numbers of which are shown within the rectangle. Such a divider 82 goes unstable when the denominator input is negative and accordingly, it is necessary to take steps to invert the polarity of the denominator signal when it goes negative and then reinsert the correct polarity afterwards.
Ln order to do this the signal Y is applied -to an inverter 84 based on an operational amp]ifier 86, for example an a~nplifier type 741. r~he signal Y is applied to the inverting input of the amplifier 86 by way of a 4K7 resis-tor 88. Since the overall gain is unity then a 41C7 resistor 90 is used to apply negative feedback.
he non-inverting input of the amplifier 86 is biased using a potentiometer 92, the bias being set at on or about zero voltsO
Analogue switches 94, 96, for example HEF 4066, are actuated as appropriate to provide a positive signal ¦Y¦ at the non-inverting -.nput of a buffer amplifier 98, for example an operational amplifier type 741, the output of which is applie~ as an input to pin 7 of the divider 82, the signal X being supplied to the pin 1, The quotient ~/IYI on pin 2 of the divider 82 has its polarity adjusted by means of another inver-ter 100 which is identic~l to thc inverter 84. ~election of the quotient sig~al X/IYI or the inverter quotient signal -X/IYI tO restore the correct polarity is done by analogue switches 102, 104. 'rhe outputs of these switches 102, 104 are applied to a buffer amplifier 106, for example an amplifier type 741~ from the output of which the desired signal ~ i& obtained.
rrhe analogue switches 94 and 102 are controlled by the output of a compara-tor 108, type LM ~9, the non-inverting input of which receives the signal Y and the inverting input of which is biased by the potentiometer 92. Irhe analogue swi-tches 96 and 104 are . . .
... . , _ . . _ ... . . . . , _ .. _ _ _ . . .
_ _, . . _ _ . , . _ _ _ _ _ _ . ., . _ _ . -- -- -- -- -- -- -- r-- ~ ~ ---- ~-- -- -- ~ -- ---- -- -- ~ ---- -- -- -- -- -- -- -- -- -- -- --1 ~.9.81 ~ PI~B 32760 controlled by the output of a comparator 110, type IJM ~9, the invertin~r input of which receives the signal Y and the non-lnverting input of which i~ biased by the potentiometer 92. By using the comparators 108, 110 the operation of the switches 94, 959 102, 104 is speeded up because an inverter is not then required.
In order to try and ensure that the input d~c. levels are ~ero it may be necessary to provide a number of potentiometers 112, 114 ar.d 116 for adjustments of the integrated circuit comprisi-llg the multiplier 82. The amplitude of the quotient signal is adjustable by means of a gain control 118~
~ ividing by zero can be problematic and can be avoided by disconnecting the output of the divider 82 for small amplitude inputs. This can be done in a variety of ways for example the inclusion of threshold circuits and gating of the output during divider denominator input ~ero crossings~
Although not shown in Figures 1 to ~, automatic gain control(s) may be inserted in one or both signal paths.
The filters 26, 28 should be high order, good rejection filters in order not to destroy the effect of fi]tering which could lead to a radio receiver inc]uding the described and illustrated demodulator receiving an adjacent channel.
If it is desired to carry out the division X/Y digitally then it will be necessary to include analogue-to-digital converters to convert the analogue signals X and Y to digital form and a digital-to-analogue converter to restore the quotient to analogue form.
In a non-illustrated modification of the invention the local oscillator 16 and the phase shifter 18 may be replaced by separate servo-controlled local oscillators with a phase difference of 90 between their outputs~
~ urther the analogue divider may comprise a multiplier in a feedbacl~ arrangement.
~ djacent channel interference c~n be removed by connecting a further low pass filter in the signal path from each of the first and second mixers 12, 14~ The rejection of adjacent signals 1~.9~81 10 PIIB 32750 when the wanted si~nal i5 small may be improved by making the, _ . . . .
numerator filter 25 of higher order than the denominator filter 28~ In this case an additional all-pass section should be incorporated into the denominator filter 28 so that the phase relationship between the two signal paths is preserved.
In the case of a data transmission system where it is important to resolve the sign of ~ ~ , only the demodulator is required to maintain a gO relative phase shift between the low frequency outputs of the first and second mixers 12, 14 and accordingly provided that this relationship is maintained the precise arrangement for achieving it is not important.
Thus a phase shift of 0 can be provided in one signal path and a phase shift of (0 ~ ~ /2) can be provided in the other signal path.
If desired instead of applying quadrature components of the local oscillator si~lals to the mixers 12, 14, the phase OL the input signal applied to one of these mixers can be shifted by 90 and the local oscillator output applied to the mixers 12, 14 directly. --, . .
4.9~81 1 PHB 3~7~o "A DIRECT MODULATION FM DEMODULATOR"
- The present invention relates to a direct modulation (frequency modulatioIl) demodulator. More particularly, the present invention relates to an F~I demodulator in which the use of inductors for filtering are avoided and the output signal is substantially independent of fading which is particularly useful in mobile receiver applications in which the strength of an input signal c&n vary quite rapidlyO
It is known to demodulate frequency-modulated signals by a method which involves sampling in some way. However, a disadvantage of this method is that when the signal has been mixed down to audio frequenciesg the s&mpling produces harmonics which lie in the audio band and cause distortion.
There also exist a number of proposals for demodulating FM
signals which do not involve sampling. British Patent Specification 1,530,602 discloses a demod~Llator comprising first and second parallel arranged signal paths whose inputs are connected to receive an incorning FM signal and whose outputs are connected to a difference amplifier. Each of the first and second si~nal paths comprises a first mixer having output connected to a low pass filter whose output is in turn connected via an automatic level control (ALC) amplifier to one input of a second mixer the output of which is connected to a respective input of the difference &mplifier~ A first signal input to the first mixer in each pa-th is the incoming FM
signal. A second signal input of the first mixer in the second path is a reference signal and the second signal input of the first mixer in the first path is the same reference signal which has been phase shifted by 9O. The outputs of the fir~t mixers are the sum and difference components of mixing. The low pass filter in each path passes the difference components, namely a cos ~ ~ t in the first path and a sin ~ ~ t in the second path, where 2a is the ampli-tude .~, 4.9~81 2 PI~B 32760 of the input signal, ~ ~ i6 the angular frequency differenee component and the amplitude of a signal from a loeal oseillator is unity. The ALC amplifiers enable the signals processed thereafter to have a st~ldard level of signal with which to work and enable the receiver to have a large working ranoee signal.
The output signals from the ALC amplifiers are supplied to a first signal input of the second mixer in its respective signal path and to respec-tive differentiators which differentiate their respective signals to give terms with arnplitude/frequeney slope and a quadrature phase shift, the output of the differentiator whose input is connected to the first path i8 connected as a second signal input to the second mixer in the second path, conversely the output of the other differentiator is connected as a second signal input to the second mixer of the first path.
The signals applied to the inputs of the difference amplifier comprise a2~ ~ cos2 S ~ t from the first path and -a2 ~ ~ sin ~ ~ t from the second path and the output from the difference &mplifier is a ~ ~ . Since this output comprises a power term, namely a2, then the output is dependent on the amplitude of the input signal which may vary considerably and rapidly if the demodulator is part of a mobile receiver.
Also known is a sine cosine frequency traeker which is disclosed in British Patent Specification 1,363,396. The frequency tracker includes a frequency discriminator which is very similar to the demodulator of British Patent Specification 1,530,602 but in which there are no ALC amplifiers. Iihe output from a difference amplifier is A2 ~ where A is the amplitude of the input signal and ~ is the difference signal between the incoming signal and a reference derived from a sweep oscilla-torO In the case of the tracking system the A2 ~ term is required in order to provide a control signal for the sweep oseillator.
United States Patent Specification 3,s68,o67 also disclo~ies a frequency diseriminator whieh is very similar to that disclosed in British Patent Specification 1,363$396~ ~his too produces an - - - -4.9~8~ 3 PHB ~,2760 i _ output which is amplitude dependent.
An object of -the present invention is to provide an FM
modulator in which amplitude and sinusoidal variations are reduced or eliminated.
According to the present invention there is provided a direct modulation FM demodulator in which an input signal ~nd a local oscillator signal are mixed in two mixers to provide outputs which are in quadra-ture phase relationship with each other, a low frequency output of one of the two mixers is phase shifted by 90 more than the phase shift in a low frequency output of the other of the two mixers to render the two signals in phase or antiphase with each other and in which the two signals are applied to a divider to obtain as an output a modulating signal contained in the input signal.
~ he phase shifting of one of the first and second mixing operations may be accomplished by di ferentiating or integrating the signal from said one mixer or by using a pair of phase shifting networks, one in each channel if only the sign of S ~
is to be determined or resolved, for example as may be the case in data transmission.
An advantage of the demodulator in accordance with the present invention is that it enables demodula-tion to be carried out at baseband due to its continuity (that i9 because there is no sampling) and it is resistant to th0 ef~ects of fading.
! The quadrature mixing of the input signal and the local oscillator signal is enabled by shifting the phase of one or another of these sig~nals by 90 prior to its being applied to ona of the two mixers.
An embodiment of a demodulator made in accordance with the present invention comprises a signal input, first and second mixers each having first and second inputs, said first inputs being connected to the signal input1 means producing a first local oscillator signal which is supplied to the second input of -the first mixer, means produc;ng a second local oscillator , . . .
,, . _ .. . . _ . _ _ _ . . . . . . . ... . .
; 4.9.81 4 PH~ 32760 I
_ _ signal which is of the same frequency as, and shifted in pha6e -- ---- by 90 relative to, the first local oscillator signal and supplyin~ it to the second input of the second mixer7 a divider having first and second inputs and an output constituting a 6 si~nal output, the first input being cornected to receive a low frequency output of one of the first and second mixers1 and the second input being connected to receive a low frequency output of the other of the first and second mixers which has been shifted in phase by 90 relative to the phase of the signal applied to the first input~ thus making the applied signals in phase or in antiphase.
The divider conveniently comprises a multiplier in a feedback arrangement. In the case of an analogue multiplier means are provided for inverting the polarity of the input signal forming the denominator in order to keep it positive~
In an embodiment of the present invention a further low pass filter is connected in the signal path from each of the first and second mixers to remove adjacent channel :interference.
If desired automatic galn control means are provided in at least one signal path from one of the first and second mixers.
Fmbodiments of the present invention will now be described, by way of example, with reference to the accompanying drawings, wherein:
Figure 1 is a block schematic circuit diagram of one embodiment of the present invention7 Figure 2 is a block schematic circuit diagram of another embodiment of the present invention7 and Figure ~ is a schematic circuit diagram of an embodiment of the type shown in ~igure 1~
Referring to Figure 1 an input FM signal 2a sin ( ~ ~ o ~ )t is applied to a first signal input of rirst and second mixers 12, 14, respectively, via an input terminal 10~ ~ signal generator or lccal oscillator 16 produces a signal, b sin ~ t, having a frequency corresponding to the carrier wave of the input signal~
This local oscillator signal is applied to a second signal input . , , , _ . . _ . _ _ _ ., _ , _ . . . _ . , .. . . , . , _ . ., 4.9.~1 5 PHB 32760 of the second mixer 14, and is applied to a phase shifter 18 which . .
shifts the phase of this signal by 90 A The phase shifted si~nal is applied to a second input of thc first mixer 12. The mixers 12, 14 which may include low pass filters produce outputs at base band frequency, in the case of the first mixer 12 the output can be represented by ab sin ~ ~ t and in the case of the second mixer 14 the output can be represented by ab cos ~ ~ t Both these signals will contain amplitude and sinusoidal variations.
These variations can be reduced and/or eliminated using a divider 22. In order to do this the phase of one of the signals has to be shifted b~ 90 in order to render it in phase or antiphase with the other of the signals. In Figure 1 this is done by applying the output, ab sin ~ ~ t, of the first mixer 12 to a differentiation circuit 20, the output of which, ~ ~ . ab cos ~ ~ t, is applied to an input X of the divider 22. A second input ~
of the divider receives the output, ab cos ~ ~'t, from the second mixer 14. The divider carries out the operation X/Y (iOeO
. ab cos ~ ~ t/ab cos ~ ~ t) to give an output signal ~ ~ on the output Z of the divider 22. The output signal ~ ~J
corresponds to the modulation component of the input iignal applied to the terminal 10 c~nd is independent of ~nplitude a Consequently the effects of fading on the output signal are reduced.
The embodiment shown in Figure 2 differs from the embodiment of Figure 1, by the fact the connections from the mixers 12, 14 to the lnputs X, Y, of the divider 22 are different. The input X of the divider 22 comprises the output ab sin ~ t from the mixer 12. The output ab cos ~ ~ t o-f the mixer 14 is integrated in an integrator 24 to provide a ~0 phase shift having an output (1/ S ~ ).ab sin ~ ~ to This comprises the input Y to the divider.
The output Z of the divider 22 again comprises ~ ~ .
In the embodiments illustrated in Figrures 1 and 2 if the circuit components in the signal paths do not proride low pass filtering to select the difference products of mixing then separate low pass filters 26, 28, shown in broken lines, should be coupled to the outputs of the mi~ers 12, 14.
.... .. . .... . .. . _ . .. .
4 9, ~1 6 PHB 32760 Referring now to Fibure 3 which i5 a schematic circuit diabram of an embodiment of the type shown in Figure 1 which uses a differer.tiator. For the sake of consistency and ease of explanation the main elements of the circuit have been enclosed in broken lines and identified by the same reference numerals as used in Figure 1. Two low pass filters 26 and 28 are connected between the mixer 12 and the differentiator 20 and between the mixer 14 and the divider 22, respectively, These filters Z6, 28 serve to filter out the high frequency lD image component and other unwanted signals in the outputs of the mixers 12 and 14.
In the following description since the mixers 12~ 14 and low pass filters 26, 28 are identical only the mixer 12 and the filter 26 will be described in detail~ The mixer 12 comprises a multiplier type SL 640C integrated circuit 30, -the pin numbers of which are shown within the rectangle. The input terminal 10 is connected to the pin 7 by way of a capacitor 32 having a value 2n7 farads. The output of the phase shifter 18 is connected to a pin 3 by way of a capacitor 34 haring a ~ value of 4r.7 farads. The pins 3 and 7 are biased ~rom a +6 volt rail by means of 10KJ-J potentiometers 36, 38 and 330K ~
resistors 40, 42, respectively. The signal output of the mixer 12 is derived from the pin 5 of the integrated circuit 300 The other connections of the integrated circuit 30 are as sho~m and the ralue of the capacitor between pin 2 and ground is 10 uF.
The low pass filter 26 is based on an operational amplifier 44, for example amplifier type 531, having negative feedback via 27K J~ resistor 46. The relevant pins of the type 531 amplifier are shown within the triangular symbol representing the amplifier.
A capacitor of 47 pF is cornected between a pin 8 o~ the amplifier 44 and its output. The output, pin 6, of the mixer 12 is connected to the inverting input, pin 2, of the amplifier 44 via an R~C.
filter network comprising series connected resistors 48 (4K7~ and 50 (2K7) and a capacitor 52 (3n3F) connected between a junction of the resistors 48~ 50 and ground. The non-inrerting input, pin .. . .. .. . .. .. . .
4;9.81 ~ P~rB ~2760 3, of the ~nplifier is bia~ed using a potentiometer 54 (4K7) and a series resistor 56 (10K) connected between the 6 volt rail and ground .
The differentiator 20 is based on ano-ther operational amplifier 58, for example amplifier -type 531. As is customary there is a feedback resistor 60 (5K6) connected between the amplifier output, pin 6, and the inverting input~ pin 29 of the amplifier 58, which input is connected to the output of the filter 26 by way of a capacitor 62 (10 ~). A non-inverting input, pin 3, of the amplifier 58 is connected to ground~ A
capacitor of 220 pF is connected between pin 8 on the amplifier 58 and its output.
As the oscillator 16 is a standard crystal oscillator circuit based on an npn transistor 64 type BF 494 it will not be described further. The output of the oscillator is connected -to a phase splitting arrangement comprising an npn transistor 66~ type ~F
494. As such an arrangement is well-known it will not be described further. The non-inverted sign~l is tapped-off from the emitter circuit of the transistor 66 and is applied by way of a capacitor 68 (4n7) to pin 3 of the integrated circuit in the mixer 14. The inverted signal is derived from the collector of the transistor 66 and applied to the phase shifter 18 which ad~u~ts the phase o~
the signal by 90.
The phase shifter 18 comprises two npn transistors 70, 72, type BF 494, which are operated as emitter followersO The inverted signal from the phase splitting arrangement is applied to the base electrode of the trarsistor 70 by way of an R.C. phase balancin~
network comprising a series variable resistor 74 (470 f~) and a capacitor 76 (100 p~) coupled to ground. An R.C. filter 78 couples the emitter o~ the transistor 70 to the base of the transistoI 72~ The 90 phase shifted signal is derived from the tap of a potentiometer 80 in the emitter circuit of the tr~lsistor 72 and is applied via the capacitor 34 to pln 3 of the integrated circuit ~Or The potentiometer 80 provides amplitude balance to the derived signal~
.. . . , , ... _ . . , _ .. _ . . .... _ . . . _ . . . .
5~j 4.9.81 8 Pl-IB 327~0 ------- The output of the differentiator 20 comprises the signal -~~~-X which is applied a& one input to the divider 22 and the output of the low pass ilter 28 comprises the signal Y which is applied as another input to the divider 22. In the illustrated embodiment the divider 22 is based on an analogue divider 82, for example an Analog Devices AD 53~D, the pin numbers of which are shown within the rectangle. Such a divider 82 goes unstable when the denominator input is negative and accordingly, it is necessary to take steps to invert the polarity of the denominator signal when it goes negative and then reinsert the correct polarity afterwards.
Ln order to do this the signal Y is applied -to an inverter 84 based on an operational amp]ifier 86, for example an a~nplifier type 741. r~he signal Y is applied to the inverting input of the amplifier 86 by way of a 4K7 resis-tor 88. Since the overall gain is unity then a 41C7 resistor 90 is used to apply negative feedback.
he non-inverting input of the amplifier 86 is biased using a potentiometer 92, the bias being set at on or about zero voltsO
Analogue switches 94, 96, for example HEF 4066, are actuated as appropriate to provide a positive signal ¦Y¦ at the non-inverting -.nput of a buffer amplifier 98, for example an operational amplifier type 741, the output of which is applie~ as an input to pin 7 of the divider 82, the signal X being supplied to the pin 1, The quotient ~/IYI on pin 2 of the divider 82 has its polarity adjusted by means of another inver-ter 100 which is identic~l to thc inverter 84. ~election of the quotient sig~al X/IYI or the inverter quotient signal -X/IYI tO restore the correct polarity is done by analogue switches 102, 104. 'rhe outputs of these switches 102, 104 are applied to a buffer amplifier 106, for example an amplifier type 741~ from the output of which the desired signal ~ i& obtained.
rrhe analogue switches 94 and 102 are controlled by the output of a compara-tor 108, type LM ~9, the non-inverting input of which receives the signal Y and the inverting input of which is biased by the potentiometer 92. Irhe analogue swi-tches 96 and 104 are . . .
... . , _ . . _ ... . . . . , _ .. _ _ _ . . .
_ _, . . _ _ . , . _ _ _ _ _ _ . ., . _ _ . -- -- -- -- -- -- -- r-- ~ ~ ---- ~-- -- -- ~ -- ---- -- -- ~ ---- -- -- -- -- -- -- -- -- -- -- --1 ~.9.81 ~ PI~B 32760 controlled by the output of a comparator 110, type IJM ~9, the invertin~r input of which receives the signal Y and the non-lnverting input of which i~ biased by the potentiometer 92. By using the comparators 108, 110 the operation of the switches 94, 959 102, 104 is speeded up because an inverter is not then required.
In order to try and ensure that the input d~c. levels are ~ero it may be necessary to provide a number of potentiometers 112, 114 ar.d 116 for adjustments of the integrated circuit comprisi-llg the multiplier 82. The amplitude of the quotient signal is adjustable by means of a gain control 118~
~ ividing by zero can be problematic and can be avoided by disconnecting the output of the divider 82 for small amplitude inputs. This can be done in a variety of ways for example the inclusion of threshold circuits and gating of the output during divider denominator input ~ero crossings~
Although not shown in Figures 1 to ~, automatic gain control(s) may be inserted in one or both signal paths.
The filters 26, 28 should be high order, good rejection filters in order not to destroy the effect of fi]tering which could lead to a radio receiver inc]uding the described and illustrated demodulator receiving an adjacent channel.
If it is desired to carry out the division X/Y digitally then it will be necessary to include analogue-to-digital converters to convert the analogue signals X and Y to digital form and a digital-to-analogue converter to restore the quotient to analogue form.
In a non-illustrated modification of the invention the local oscillator 16 and the phase shifter 18 may be replaced by separate servo-controlled local oscillators with a phase difference of 90 between their outputs~
~ urther the analogue divider may comprise a multiplier in a feedbacl~ arrangement.
~ djacent channel interference c~n be removed by connecting a further low pass filter in the signal path from each of the first and second mixers 12, 14~ The rejection of adjacent signals 1~.9~81 10 PIIB 32750 when the wanted si~nal i5 small may be improved by making the, _ . . . .
numerator filter 25 of higher order than the denominator filter 28~ In this case an additional all-pass section should be incorporated into the denominator filter 28 so that the phase relationship between the two signal paths is preserved.
In the case of a data transmission system where it is important to resolve the sign of ~ ~ , only the demodulator is required to maintain a gO relative phase shift between the low frequency outputs of the first and second mixers 12, 14 and accordingly provided that this relationship is maintained the precise arrangement for achieving it is not important.
Thus a phase shift of 0 can be provided in one signal path and a phase shift of (0 ~ ~ /2) can be provided in the other signal path.
If desired instead of applying quadrature components of the local oscillator si~lals to the mixers 12, 14, the phase OL the input signal applied to one of these mixers can be shifted by 90 and the local oscillator output applied to the mixers 12, 14 directly. --, . .
Claims (7)
PROPERTY OR PRIVILEGE IS CLAIMED ARE DEFINED AS FOLLOWS:
1. A direct modulation FM demodulator characterised in that an input signal and a local oscillator signal are mixed in two mixers to provide outputs which are in a quadrature phase relationship with each other, in that an integrator is coupled to a low frequency output of one of the two mixers to shift the phase of said output by 90°
more than the phase shift in a low frequency output of the other of the two mixers thereby providing two signals which are in phase or antiphase with each other, and in that the two signals are applied to a divider to obtain as an output a modulating signal contained in the input signal.
more than the phase shift in a low frequency output of the other of the two mixers thereby providing two signals which are in phase or antiphase with each other, and in that the two signals are applied to a divider to obtain as an output a modulating signal contained in the input signal.
2. A demodulator as claimed in Claim 1, character-ised in that the phase of the input signal applied to one of the two mixers is shifted by 90° relative to the phase of the input signal applied to the other of the two mixers, and in that the local oscillator signal applied to the two mixers has substantially the same phase.
3. A demodulator as claimed in Claim 1, character-ised in that the two mixers each have first and second inputs, said first inputs being connected to receive the input signal, in that means are provided for producing a first local oscillator signal which is supplied to the second input of one of the two mixers, in that means are provided for producing a second local oscillator signal which is of the same frequency as, and is shifted in phase by 90° relative to, the first local oscillator signal and supplying it to the second input of the other of the two mixers, and in that the divider has first and second inputs and an output constituting a signal output, the first input being connected to receive the output of the inte-grator and the second input being connected to receive a low frequency output of the other of the two mixers.
4. A demodulator as claimed in Claim 3, character-ised in that the divider comprises a multiplier in a feed back arrangement.
5. A demodulator as claimed in Claim 4, character-ised in that means are provided for inverting the polarity of the denominator input signal to the divider and the output signal of the divider as required.
6. A demodulator as claimed in Claim 1, character-ised in that a low pass filter is connected in the signal path from each of the first and second mixers to remove adjacent channel interference.
7. A demodulator as claimed in Claim 6, character-ised in that the low pass filter for the signal serving as the numerator in the divider is of higher order than the low pass filter for the signal serving as the denom-inator in the divider, and in that the low pass filter for the signal serving as the denominator in the divider, includes an all-pass section.
Applications Claiming Priority (2)
| Application Number | Priority Date | Filing Date | Title |
|---|---|---|---|
| GB8105456A GB2094079A (en) | 1981-02-20 | 1981-02-20 | Fm demodulator |
| GB8105456 | 1981-02-20 |
Publications (1)
| Publication Number | Publication Date |
|---|---|
| CA1184256A true CA1184256A (en) | 1985-03-19 |
Family
ID=10519858
Family Applications (1)
| Application Number | Title | Priority Date | Filing Date |
|---|---|---|---|
| CA000396534A Expired CA1184256A (en) | 1981-02-20 | 1982-02-18 | Direct modulation fm demodulator |
Country Status (8)
| Country | Link |
|---|---|
| US (1) | US4488119A (en) |
| EP (1) | EP0059000B1 (en) |
| JP (1) | JPS57152708A (en) |
| AU (1) | AU548502B2 (en) |
| CA (1) | CA1184256A (en) |
| DE (1) | DE3265319D1 (en) |
| DK (1) | DK160029C (en) |
| GB (1) | GB2094079A (en) |
Families Citing this family (30)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| GB2106359B (en) * | 1981-09-24 | 1985-07-03 | Standard Telephones Cables Ltd | Direct conversion radio receiver for fm signals |
| FR2538634B1 (en) * | 1982-12-24 | 1985-07-19 | Int Standard Electric Corp | DIRECT CONVERSION RADIO RECEIVER FOR FREQUENCY MODULATED SIGNALS |
| AT398148B (en) * | 1984-07-25 | 1994-09-26 | Sat Systeme Automatisierung | Method for the demodulation of a frequency-modulated signal, and circuit arrangement for carrying out the method |
| GB8528541D0 (en) * | 1985-11-20 | 1985-12-24 | Devon County Council | Fm demodulator |
| GB2184304B (en) * | 1985-12-12 | 1989-10-11 | Rank Taylor Hobson Ltd | Velocity measuring apparatus |
| US4980687A (en) * | 1988-10-13 | 1990-12-25 | Systron Donner | Digital demodulator |
| JP2953365B2 (en) * | 1995-11-17 | 1999-09-27 | 日本電気株式会社 | Quadrature demodulator |
| US6314279B1 (en) * | 1998-06-29 | 2001-11-06 | Philips Electronics North America Corporation | Frequency offset image rejection |
| US6061551A (en) | 1998-10-21 | 2000-05-09 | Parkervision, Inc. | Method and system for down-converting electromagnetic signals |
| US7515896B1 (en) | 1998-10-21 | 2009-04-07 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same, and aperture relationships |
| US6370371B1 (en) | 1998-10-21 | 2002-04-09 | Parkervision, Inc. | Applications of universal frequency translation |
| US7039372B1 (en) * | 1998-10-21 | 2006-05-02 | Parkervision, Inc. | Method and system for frequency up-conversion with modulation embodiments |
| US6813485B2 (en) | 1998-10-21 | 2004-11-02 | Parkervision, Inc. | Method and system for down-converting and up-converting an electromagnetic signal, and transforms for same |
| US7236754B2 (en) | 1999-08-23 | 2007-06-26 | Parkervision, Inc. | Method and system for frequency up-conversion |
| US7209725B1 (en) | 1999-01-22 | 2007-04-24 | Parkervision, Inc | Analog zero if FM decoder and embodiments thereof, such as the family radio service |
| US6853690B1 (en) | 1999-04-16 | 2005-02-08 | Parkervision, Inc. | Method, system and apparatus for balanced frequency up-conversion of a baseband signal and 4-phase receiver and transceiver embodiments |
| US6879817B1 (en) | 1999-04-16 | 2005-04-12 | Parkervision, Inc. | DC offset, re-radiation, and I/Q solutions using universal frequency translation technology |
| US7065162B1 (en) | 1999-04-16 | 2006-06-20 | Parkervision, Inc. | Method and system for down-converting an electromagnetic signal, and transforms for same |
| US7693230B2 (en) | 1999-04-16 | 2010-04-06 | Parkervision, Inc. | Apparatus and method of differential IQ frequency up-conversion |
| US7110444B1 (en) | 1999-08-04 | 2006-09-19 | Parkervision, Inc. | Wireless local area network (WLAN) using universal frequency translation technology including multi-phase embodiments and circuit implementations |
| US8295406B1 (en) | 1999-08-04 | 2012-10-23 | Parkervision, Inc. | Universal platform module for a plurality of communication protocols |
| US7010286B2 (en) | 2000-04-14 | 2006-03-07 | Parkervision, Inc. | Apparatus, system, and method for down-converting and up-converting electromagnetic signals |
| US7454453B2 (en) | 2000-11-14 | 2008-11-18 | Parkervision, Inc. | Methods, systems, and computer program products for parallel correlation and applications thereof |
| US6459333B1 (en) * | 2001-02-05 | 2002-10-01 | Motorola, Inc. | Differentiate and divide FM demodulator |
| US6943847B2 (en) * | 2001-08-31 | 2005-09-13 | Texas Instruments Incorporated | FM demodulator for SECAM decoder |
| US7072427B2 (en) | 2001-11-09 | 2006-07-04 | Parkervision, Inc. | Method and apparatus for reducing DC offsets in a communication system |
| US7460584B2 (en) | 2002-07-18 | 2008-12-02 | Parkervision, Inc. | Networking methods and systems |
| US7379883B2 (en) | 2002-07-18 | 2008-05-27 | Parkervision, Inc. | Networking methods and systems |
| CN1988397B (en) * | 2005-12-21 | 2011-05-25 | 上海贝岭股份有限公司 | Frequency modulation receiver and its demodulation method |
| US7843627B2 (en) * | 2008-11-26 | 2010-11-30 | Agilent Technologies, Inc. | Coherent demodulation with reduced latency adapted for use in scanning probe microscopes |
Family Cites Families (15)
| Publication number | Priority date | Publication date | Assignee | Title |
|---|---|---|---|---|
| US3101448A (en) * | 1954-12-23 | 1963-08-20 | Gen Electric | Synchronous detector system |
| US3035231A (en) * | 1959-01-16 | 1962-05-15 | Gen Electric | Frequency difference discriminator |
| US3250904A (en) * | 1961-09-27 | 1966-05-10 | Vasu George | Phase-shift computer and control system |
| US3609555A (en) * | 1967-07-19 | 1971-09-28 | Ibm | Digital fm receiver |
| US3500217A (en) * | 1967-07-31 | 1970-03-10 | Us Navy | Frequency discriminator employing quadrature demodulation techniques |
| US3568067A (en) * | 1969-06-13 | 1971-03-02 | Collins Radio Co | Frequency discriminator with output indicative of difference between input and local reference signals |
| US3569853A (en) * | 1969-07-01 | 1971-03-09 | Communications Satellite Corp | Phase-lock loop with tangent function phase comparator |
| US3748590A (en) * | 1972-04-14 | 1973-07-24 | Singer Co | Sine cosine frequency tracker |
| GB1556045A (en) * | 1975-07-02 | 1979-11-21 | Marconi Co Ltd | Analogue phase sensitive detectors |
| GB1530602A (en) * | 1975-10-14 | 1978-11-01 | Standard Telephones Cables Ltd | Demodulator for fm signals |
| HU175236B (en) * | 1977-01-10 | 1980-06-28 | Hiradastech Ipari Kutato | Method and apparaus to receive and generate frquency modulated signals |
| JPS5389354A (en) * | 1977-01-18 | 1978-08-05 | Toshiba Corp | Digital synchronous detector |
| US4253067A (en) * | 1978-12-11 | 1981-02-24 | Rockwell International Corporation | Baseband differentially phase encoded radio signal detector |
| US4270221A (en) * | 1979-10-17 | 1981-05-26 | Rca Corporation | Phaselocked receiver with orderwire channel |
| US4525862A (en) * | 1980-07-02 | 1985-06-25 | Motorola, Inc. | Transform modulation system |
-
1981
- 1981-02-20 GB GB8105456A patent/GB2094079A/en not_active Withdrawn
-
1982
- 1982-01-29 EP EP82200106A patent/EP0059000B1/en not_active Expired
- 1982-01-29 DE DE8282200106T patent/DE3265319D1/en not_active Expired
- 1982-02-04 US US06/345,863 patent/US4488119A/en not_active Expired - Fee Related
- 1982-02-17 DK DK069482A patent/DK160029C/en not_active IP Right Cessation
- 1982-02-17 JP JP57022929A patent/JPS57152708A/en active Granted
- 1982-02-18 CA CA000396534A patent/CA1184256A/en not_active Expired
- 1982-02-19 AU AU80615/82A patent/AU548502B2/en not_active Ceased
Also Published As
| Publication number | Publication date |
|---|---|
| JPH0237721B2 (en) | 1990-08-27 |
| EP0059000B1 (en) | 1985-08-14 |
| DK160029C (en) | 1991-06-03 |
| DK160029B (en) | 1991-01-14 |
| GB2094079A (en) | 1982-09-08 |
| AU548502B2 (en) | 1985-12-12 |
| AU8061582A (en) | 1982-08-26 |
| DE3265319D1 (en) | 1985-09-19 |
| US4488119A (en) | 1984-12-11 |
| DK69482A (en) | 1982-08-21 |
| EP0059000A1 (en) | 1982-09-01 |
| JPS57152708A (en) | 1982-09-21 |
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